Communication system

ABSTRACT

At the transmitter side, carrier waves are modulated according to an input signal for producing relevant signal points in a signal space diagram. The input signal is divided into, two, first and second, data streams. The signal points are divided into signal point groups to which data of the first data stream are assigned. Also, data of the second data stream are assigned to the signal points of each signal point group. A difference in the transmission error rate between first and second data streams is developed by shifting the signal points to other positions in the space diagram expressed at least in the polar coordinate system. At the receiver side, the first and/or second data streams can be reconstructed from a received signal. In TV broadcast service, a TV signal is divided by a transmitter into low and high frequency band components which are designated as first and second data streams respectively. Upon receiving the TV signal, a receiver can reproduce only the low frequency band component or both the low and high frequency band components, depending on its capability. Furthermore, a communication system based on an OFDM system is utilized for data transmission of a plurality of subchannels, wherein the subchannels are differentiated by changing the length of a guard time slot or a carrier wave interval of a symbol transmission time slot, or changing the transmission electric power of the carrier.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation-in-part of application Ser. No.07/857,627, filed Mar. 25, 1992, pending.This is a reissue applicationof U.S. Pat. No. 5,600,672, issued Feb. 4, 1997, and a divisionalapplication of reissue application Ser. No. 09/244,037, filed Feb. 4,1999, which is also a reissue application of U.S. Pat. No. 5,600,672,issued Feb. 4, 1997 which is a Continuation-In-Part of application Ser.No. 07/857,627, filed Mar. 25, 1992 now abandoned. Further reissuedivisional applications have been filed, all of which are reissues ofU.S. Pat. No. 5,600,672. These further applications are: Ser. No.09/677,421, filed Oct. 5, 2000; Ser. No. 09/678,014, filed Oct. 5, 2000;Ser. No. 09/677,420, filed Oct. 5, 2000; Ser. No. 09/680,177, filed Oct.5, 2000; Ser. No. 09/680,176, filed Oct. 5, 2000; Ser. No. 09/686,467,filed Oct. 12, 2000; Ser. No. 09/686,463, filed Oct. 12, 2000; Ser. No.09/686,466, filed Oct. 12, 2000; Ser. No. 09/688,028, filed Oct. 12,2000; Ser. No. 09/686,464, filed Oct. 12, 2000; Ser. No. 09/686,465,filed Oct. 12, 2000; Ser. No. 09/666,012, filed Sep. 19, 2000; Ser. No.09/667,525, filed Sep. 21, 2000; Ser. No. 09/667,438, filed Sep. 21,2000; Ser. No. 09/668,068, filed Sep. 25, 2000; Ser. No. 09/669,916,filed Sep. 25, 2000; Ser. No. 09/672,948, filed Sep. 29, 2000; Ser. No.09/672,946, filed Sep. 29, 2000; Ser. No. 09/672,947, filed Sep. 29,2000; Ser. No. 10/133,347, filed Apr. 29, 2002; Ser. No. 10/133,364,filed Apr. 29, 2002; Ser. No. 10/692,469, filed Oct. 24, 2003; Ser. No.10/693,526, filed Oct. 27, 2003; Ser. No. 10/635,468, filed Aug. 7,2003; Ser. No. 10/690,297, filed Oct. 27, 2003; Ser. No. 10/860,666,filed Jun. 4, 2004; Ser. No. 10/782,411, filed Feb. 20, 2004; Ser. No.10/783,588, filed Feb. 23, 2004; Ser. No. 10/773,811, filed Feb. 9,2004; Ser. No. 10/882,126, filed Jun. 30, 2004; Ser. No. 10/885,572,filed Jul. 7, 2004; Ser. No. 10/911,680, filed Aug. 5, 2004; and Ser.No. 11/038,006, filed Jan. 19, 2006.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a communication system for transmissionand reception of a digital signal through modulation of its carrier waveand demodulation of the modulated signal.

2. Description of the Prior Art

Digital signal communication systems have been used in various fields.Particularly, digital video signal transmission techniques have beenimproved remarkably.

Among them is a digital TV signal transmission method. So far, suchdigital TV signal transmission systems are in particular use fortransmission between TV stations. They will soon be utilized forterrestrial and/or satellite broadcast service in every country of theworld.

The TV broadcast systems including HDTV, PCM music, FAX, and otherinformation services are now demanded to increase desired data inquantity and quality for satisfying millions of sophisticated viewers.In particular, the data has to be increased in a given bandwidth offrequency allocated for TV broadcast service. The data to be transmittedis always abundant and provided as much as handled with up-to-datetechniques of the time. It is ideal to modify or change the existingsignal transmission system corresponding to an increase in the dataamount with time.

However, the TV broadcast service is a public business and cannot gofurther without considering the interests and benefits of viewers. It isessential to have any new service compatible with existing TV receiversand displays. More particularly, the compatibility of a system is muchdesired for providing both old and new services simultaneously or onenew service which can be intercepted by both the existing and advancedreceivers.

It is understood that any new digital TV broadcast system to beintroduced has to be arranged for data extension in order to respond tofuture demands and technological advantages and also, for compatibilityto allow the existing receivers to receive transmissions.

The expansion capability and compatible performance of the prior artdigital TV system will be explained.

A digital satellite TV system is known in which NTSC TV signalscompressed to an about 6 Mbps are muitiplexed multiplexed by timedivision modulation of 4 PSK and transmitted on 4 to 20 channels whileHDTV signals are carried on a signal channel. Another digital HDTVsystem is provided in which HDTV video data compressed to as small as 15Mbps are transmitted on a 16 or 32 QAM signal through ground stations.

Such a known satellite system permits HDTV signals to be carried on thechannel by a conventional manner, thus occupying a band of frequenciesequivalent to the same channels of NTSC signals. This causes thecorresponding NTSC channels to be unavailable during the transmission ofthe HDTV signal. Also, the compatibility between NTSC and HDTV receiversor displays is hardly concerned and data expansion capability needed formatching a future advanced mode is utterly disregarded.

Such a common terrestrial HDTV system offers an HDTV service onconventional 16 or 32 QAM signals without any modification. In anyanalogue TV broadcast service, there are developed a lot of signalattenuating or shadow regions within its service area due to structuralobstacles, geographical inconveniences, or signal interference from aneighbor station. When the TV signal is an analogue from form, it can beintercepted more or less at such signal attenuating regions although itsreproduced picture is low in quality. If the TV signal is a digitalform, it can rarely be reproduced at an acceptable level within theregions. This disadvantage is critically hostile to the development ofany digital TV system.

SUMMARY OF THE INVENTION

It is an object of the present invention, for solving the foregoingdisadvantages, to provide a communication system arranged for compatibleuse for both the existing NTSC and newly introduced HDTV broadcastservices, particularly via satellite and also, for minimizing signalattenuating or shadow region of its service area on the grounds ground.

A communication system according to the present invention intentionallyvaries signal points, which used to be disposed at uniform intervals, toperform the signal transmission and reception. For example, if appliedto a QAM signal, the communication system comprises two major sections:a transmitter having a signal input circuit, a modulator circuit forproducing m numbers of signal points, in a signal vector field throughmodulation of a plurality of out-of-phase carrier waves using an inputsignal supplied from the input circuit, and a transmitter circuit fortransmitting a resultant modulated signal; and a receiver having aninput circuit for receiving the modulated signal, a demodulator circuitfor demodulating one-bit signal points of a QAM carrier wave, and anoutput circuit.

In operation, the input signal containing a first data stream of nvalues and a second data stream is fed to the modulator circuit of thetransmitter where a modified m-bit QAM carrier wave is producedrepresenting m signal points in a vector field. The m signal points aredivided into n signal point groups to which the n values of the firstdata stream are assigned respectively. Also, data of the second datastream are assigned to m/n signal points or sub groups of each signalpoint group. Then, a resultant transmission signal is transmitted fromthe transmitter circuit. Similarly, a third data stream can bepropagated.

At the p-bit demodulator circuit, p>m, of the receiver, the first datastream of the transmission signal if is first demodulated throughdividing p signal points in a signal space diagram into n signal pointgroups. Then, the second data stream is demodulated through assigningp/n values to p/n signal points of each corresponding signal point groupfor reconstruction of both the first and second data streams. If thereceiver is at P=n, the n signal point groups are reclaimed and assignedthe n values for demodulation and reconstruction of the first datastream.

Upon receiving the same transmission signal from the transmitter, areceiver equipped with a large sized antenna and capable of large-datamodulation can reproduce both the first and second data streams. Areceiver equipped with a small sized antenna and capable of small-datamodulation can reproduce the first data stream only. Accordingly, thecompatibility of the signal transmission system will be ensured. Whenthe first data stream is an NTSC TV signal or low frequency bandcomponent of an HDTV signal and the second data stream is a highfrequency band component of the HDTV signal, the small-data modulationreceiver can reconstruct the NTSC TV signal and the large-datamodulation receiver can reconstruct the HDTV signal. As understood, adigital NTSC/HDTV simultaneous broadcast service will be feasible usingthe compatibility of the signal transmission system of the presentinvention.

More specifically, the communication system of the present inventioncomprises: a transmitter having a signal input circuit, a modulatorcircuit for producing m signal point, points in a signal vector fieldthrough modulation of a plurality of out-of-phase carrier waves using aninput signal supplied from the input, and a transmitter circuit fortransmitting a resultant modulated signal, in which the main procedureincludes receiving an input signal containing a first data stream of nvalues and a second data stream, dividing the m signal points of thesignal into n signal point groups, assigning the n values of the firstdata stream to the n signal point groups respectively, assigning data ofthe second data stream to signal points of each signal point grouprespectively, and transmitting the resultant modulated signal; and areceiver having an input circuit for receiving the modulated signal, ademodulator circuit for demodulating p signal points of a QAM carrierwave, and an output circuit, in which the main procedure includesdividing the p signal points into n signal point groups, demodulatingthe first data stream of which n values are assigned to the n signalpoint groups respectively, and demodulating the second data stream ofwhich p/n values are assigned to p/n signal points of each signal pointgroup respectively. For example, a transmitter produces a modified m-bitQAM signal of which first, second, and third data streams, each carryingn values, are assigned to relevant signal point groups with a modulator.The signal can be intercepted and the first data stream only reproducedby a first receiver, both the first and second data streams can bereproduced by a second receiver, and all the first, second, and thirdstreams can be reproduced by a third receiver.

More particularly, a receiver capable of demodulation of n-bit data canreproduce n bits from a multiple-bit modulated carrier wave carryingm-bit data where m>n, thus allowing the communication system to havecompatibility and capability of future extension. Also, a multi-levelsignal transmission will be possible by shifting the signal points ofQAM so that a nearest signal point to the origin point of I-axis andQ-axis coordinates is spaced nf from the origin where f is the distanceof the nearest point from each axis and n is more than 1.

Accordingly, a compatible digital satellite broadcast service for boththe NTSC and HDTV systems will be feasible when the first data streamcarries an NTSC signal and the second data stream carries a differencesignal between NTSC and HDTV. Hence, the capability of corresponding toan increase in the data amount to be transmitted will be ensured. Also,on the ground, the service area will be increased while signalattenuating areas are decreased.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic view of the entire arrangement of a signaltransmission system showing a first embodiment of the present invention;

FIG. 2 is a block diagram of a transmitter of the first embodiment;

FIG. 3 is a vector diagram showing a transmission signal of the firstembodiment;

FIG. 4 is a vector diagram showing a transmission signal of the firstembodiment;

FIG. 5 is a view showing an assignment of binary codes to signal pointsaccording to the first embodiment;

FIG. 6 is a view showing an assignment of binary codes to signal pointgroups according to the first embodiment;

FIG. 7 is a view showing an assignment of binary codes to signal pointsin each signal point group according to the first embodiment;

FIG. 8 is a view showing another assignment of binary codes to signalpoint groups and their signal points according to the first embodiment;

FIG. 9 is a view showing threshold values of the signal point groupsaccording to the first embodiment;

FIG. 10 is a vector diagram of a modified 16 QAM signal of the firstembodiment;

FIG. 11 is a graphic diagram showing the relationship between antennaradius r₂ and transmission energy ratio n according to the firstembodiment;

FIG. 12 is a view showing the signal points of a modified 64 QAM signalof the first embodiment;

FIG. 13 is a graphic diagram showing the relationship between antennaradius r₃ and transmission energy ratio n according to the firstembodiment;

FIG. 14 is a vector diagram showing signal point groups and their signalpoints of the modified 64 QAM signal of the first embodiment;

FIG. 15 is an explanatory view showing the relationship between A₁ andA₂ of the modified 64 QAM signal of the first embodiment;

FIG. 16 is a graph diagram showing the relationship between antennaradius r₂ and r₃ and transmission energy ratio n₁₆ and n₆₄ respectivelyaccording to the first embodiment;

FIG. 17 is a block diagram of a digital transmitter of the firstembodiment;

FIG. 18 is a signal space diagram of a 4 PSK modulated signal of thefirst embodiment;

FIG. 19 is a block diagram of a first receiver of the first embodiment;

FIG. 20 is a signal space diagram of a 4 PSK modulated signal of thefirst embodiment;

FIG. 21 is a block diagram of a second receiver of the first embodiment;

FIG. 22 is a vector diagram of a modified 16 QAM signal of the firstembodiment;

FIG. 23 is a vector diagram of a modified 64 QAM signal of the firstembodiment;

FIG. 24 is a flowchart showing the operation of the first embodiment;

FIGS. 25(a) and 25(b) are vector diagrams respectively showing an 8 anda 16 QAM signal of the first embodiment;

FIG. 26 is a block diagram of a third receiver of the first embodiment;

FIG. 27 is a view showing signal points of the modified 64 QAM signal ofthe first embodiment;

FIG. 28 is a flowchart showing another the operation of the firstembodiment;

FIG. 29 is a schematic view of the entire arrangement of a signaltransmission system showing a third embodiment of the present invention;

FIG. 30 is a block diagram of a first video encoder of the thirdembodiment;

FIG. 31 is a block diagram of a first video decoder of the thirdembodiment;

FIG. 32 is a block diagram of a second video decoder of the thirdembodiment;

FIG. 33 is a block diagram of a third video decoder of the thirdembodiment;

FIG. 34 is an explanatory view showing a time multiplexing of D₁, D₂,and D₃ signals according to the third embodiment;

FIG. 35 is an explanatory view showing another time multiplexing of D₁,D₂, and D₃ signals according to the third embodiment;

FIG. 36 is an explanatory view showing a further time multiplexing ofD₁, D₂, and D₃ signals according to the third embodiment;

FIG. 37 is a schematic view of the entire arrangement of a signaltransmission system showing a fourth embodiment of the presentinvention;

FIG. 38 is a vector diagram of a modified 16 QAM signal of the thirdembodiment;

FIG. 39 is a vector diagram of the modified 16 QAM signal of the thirdembodiment;

FIG. 40 is a vector diagram of a modified 64 QAM signal of the thirdembodiment;

FIG. 41 is a diagram of assignment of data components on a time baseaccording to the third embodiment;

FIG. 42 is a diagram of assignment of data components on a time base inTDMA action according to the third embodiment;

FIG. 43 is a block diagram of a carrier reproducing circuit of the thirdembodiment;

FIG. 44 is a diagram showing the principle of carrier wave reproductionaccording to the third embodiment;

FIG. 45 is a block diagram of a carrier reproducing circuit for reversemodulation of the third embodiment;

FIG. 46 is a diagram showing an assignment of signal points of the 16QAM signal of the third embodiment;

FIG. 47 is a diagram showing an assignment of signal points of the 64QAM signal of the third embodiment;

FIG. 48 is a block diagram of a carrier reproducing circuit for 16×multiplication of the third embodiment;

FIG. 49 is an explanatory view showing a time multiplexing of D_(V1),D_(H1), D_(V2), D_(H2), D_(V3), and D_(H3) signals according to thethird embodiment;

FIG. 50 is an explanatory view showing a TDMA time multiplexing ofD_(V1), D_(H1), D_(V2), D_(H2), D_(V3), and D_(H3) signals according tothe third embodiment;

FIG. 51 is an explanatory view showing another TDMA time multiplexing ofthe D_(V1), D_(H1), D_(V2), D_(H2), D_(V3), and D_(H3) signals accordingto the third embodiment;

FIG. 52 is a diagram showing a signal interference region in a knowntransmission method according to the fourth embodiment;

FIG. 53 is a diagram showing signal interference regions in amulti-level signal transmission method according to the fourthembodiment;

FIG. 54 is a diagram showing signal attenuating regions in the knowntransmission method according to the fourth embodiment;

FIG. 55 is a diagram showing signal attenuating regions in themulti-level signal transmission method according to the fourthembodiment;

FIG. 56 is a diagram showing a signal interference region between twodigital TV stations according to the fourth embodiment;

FIG. 57 is a diagram showing an assignment of signal points of modified4 ASK signal of the fifth embodiment;

FIG. 58 is a diagram showing another assignment of signal points of themodified 4 ASK signal of the fifth embodiment;

FIGS. 59(a) and 59(b) are diagrams showing assignment of signal pointsof the modified 4 ASK signal of the fifth embodiment and FIGS. 59(c) and59(d) are diagrams respectively showing the slice levels of themodulated 4 ASK signal in subchannels 1 and 2;

FIG. 60 is a diagram showing another assignment of signal points of themodified 4 ASK signal of the fifth embodiment when the C/N rate is low;

FIG. 61 shows a 4- and 8-level VSB transmitter according to the fifthembodiment of the invention;

FIG. 62(a) is a wave spectrum diagram of the ASK signal, i.e., amulti-value VSB signal before filtering, in the fifth embodiment of theinvention and FIG. 62(b) is a wave spectrum diagram showing thecharacteristics of the filtered VSB signal;

FIG. 63 is a block diagram of a 4-, 8-, and 16-level VSB receiver in thefifth embodiment of the invention;

FIG. 64 is a block diagram of a video signal transmitter of the fifthembodiment;

FIG. 65 is a block diagram of a TV receiver of the fifth embodiment;

FIG. 66 is a block diagram of another TV receiver of the fifthembodiment;

FIG. 67 is a block diagram of a satellite-to-ground TV receiver of thefifth embodiment;

FIG. 68(a) is an 8-level VSB constellation map in the fifth and sixthembodiments of the invention;

FIG. 68(b) is an 8-level VSB constellation map in the fifth and sixthembodiments of the invention;

FIG. 68(c) is an 8-level VSB signal-time waveform diagram in the fifthand sixth embodiments of the invention;

FIG. 69 is a block diagram of a video encoder of the fifth embodiment;

FIG. 70 is a block diagram of a video encoder of the fifth embodimentcontaining one divider circuit;

FIG. 71 is a block diagram of a video decoder of the fifth embodiment;

FIG. 72 is a block diagram of a video decoder of the fifth embodimentcontaining one mixer circuit;

FIG. 73 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 74(a) is a block diagram of a video decoder of the fifthembodiment;

FIG. 74(b) is a diagram showing another time assignment of datacomponents of the transmission signal according to the fifth embodiment;

FIG. 75 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 76 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 77 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 78 is a block diagram of a video decoder of the fifth embodiment;

FIG. 79 is a diagram showing a time assignment of data components of athree-level transmission signal according to the fifth embodiment;

FIG. 80 is a block diagram of another video decoder of the fifthembodiment;

FIG. 81 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 82 is a block diagram of a video decoder for D₁ signal of the fifthembodiment;

FIG. 83 is a graphic diagram showing the relationship between frequencyand time of a frequency modulated signal according to the fifthembodiment;

FIG. 84 is a block diagram of a magnetic record/playback apparatus ofthe fifth embodiment;

FIG. 85 is a graphic diagram showing the relationship between C/N andlevel according to the second embodiment;

FIG. 86 is a graphic diagram showing the relationship between C/N andtransmission distance according to the second embodiment;

FIG. 87 is a block diagram of a transmissiontransmitter of the secondembodiment;

FIG. 88 is a block diagram of a receiver of the second embodiment;

FIG. 89 is a graphic diagram showing the relationship between C/N anderror rate according to the second embodiment;

FIG. 90 is a diagram showing signal attenuating regions in thethree-level transmission of the fifth embodiment;

FIG. 91 is a diagram showing signal attenuating regions in thefour-level transmission of athe sixth embodiment;

FIG. 92 is a diagram showing the four-level transmission of the sixthembodiment;

FIG. 93 is a block diagram of a divider of the sixth embodiment;

FIG. 94 is block diagram of a mixer of the sixth embodiment;

FIG. 95 is a diagram showing another four-level transmission of thesixth embodiment;

FIG. 96 is a view of signal propagation of a known digital TV broadcastsystem;

FIG. 97 is a view of signal propagation of a digital TV broadcast systemaccording to the sixth embodiment;

FIG. 98 is a diagram showing a four-level transmission of the sixthembodiment;

FIG. 99 is a vector diagram of a 16 SRQAM signal of the thirdembodiment;

FIG. 100 is a vector diagram of a 32 SRQAM signal of the thirdembodiment;

FIG. 101 is a graphic diagram showing the relationship between C/N anderror rakerate according to the third embodiment;

FIG. 102 is a graphic diagram showing the relationship between C/N anderror rate according to the third embodiment;

FIG. 103 is a graphic diagram showing the relationship between shiftdistance n and C/N needed for transmission according to the thirdembodiment;

FIG. 104 is a graphic diagram showing the relationship between shiftdistance n and C/N needed for transmission according to the thirdembodiment;

FIG. 105 is a graphic diagram showing the relationship between signallevel and distance from a transmitter antenna in terrestrial broadcastservice according to the third embodiment;

FIG. 106 is a diagram showing a service area of the 32 SRQAM signal ofthe third embodiment;

FIG. 107 is a diagram showing a service area of the 32 SRQAM signal ofthe third embodiment;

FIG. 108(a) is a diagram showing a frequency distribution profile of aconventional TV signal;

FIG. 108(b) is a diagram showing a frequency distribution profile of aconventional two-layer TV signal;

FIG. 108(c) is a diagram showing threshold values of the thirdembodiment;

FIG. 108(d) is a diagram showing a frequency distribution profile oftwo-layer OFDM carriers of the ninth embodiment, and FIG. 108(e) is adiagram showing threshold values for three-layer OFDM of the ninthembodiment;

FIG. 109 is a diagram showing a time assignment of the TV signal of thethird embodiment;

FIG. 110 is a diagram showing a principle of C-CDM of the thirdembodiment;

FIG. 111 is a view showing an assignment of codes according to the thirdembodiment;

FIG. 112 is a view showing an assignment of an extended 36 QAM accordingto the third embodiment;

FIG. 113 is a view showing a frequency assignment of a modulation signalaccording to the fifth embodiment;

FIG. 114 is a block diagram showing a magnetic recording/playbackapparatus according to the fifth embodiment;

FIG. 115 is a block diagram showing a transmitter/receiver of a portabletelephone according to the eighth embodiment;

FIG. 116 is a block diagram showing base stations according to theeighth embodiment;

FIG. 117 is a view illustrating communication capacities and trafficdistribution of a conventional system;

FIG. 118 is a view illustrating communication capacities and trafficdistribution according to the eighth embodiment;

FIG. 119(a) is a diagram showing a time slot assignment of aconventional system;

FIG. 119(b) is a diagram showing a time slot assignment according to theeighth embodiment;

FIG. 120(a) is a diagram showing a time slot assignment of aconventional TDMA system;

FIG. 120(b) is a diagram showing a time slot assignment according to aTDMA system of the eighth embodiment;

FIG. 121 is a block diagram showing a one-level transmitter/receiveraccording to the eighth embodiment;

FIG. 122 is a block diagram showing a two-level transmitter/receiveraccording to the eighth embodiment;

FIG. 123 is a block diagram showing an OFDM type transmitter/receiveraccording to the ninth embodiment;

FIG. 124 is a view illustrating a principle of the OFDM system accordingto the ninth embodiment;

FIG. 125(a) is a view showing a frequency assignment of a modulationsignal of a conventional system;

FIG. 125(b) is a view showing a frequency assignment of a modulationsignal according to the ninth embodiment;

FIG. 126(a) is a view showing a frequency assignment of an OFDM signalof the ninth embodiment, wherein no weighting is applied;

FIG. 126(b) is a view showing a frequency assignment of an OFDM signalof the ninth embodiment, wherein two channels of two-layer OFDM areweighted by transmission electric power;

FIG. 126(c) is a view showing a frequency assignment of an OFDM signalof the ninth embodiment, wherein carrier intervals are doubled byweighting;

FIG. 126(d) is a view showing a frequency assignment of an OFDM signalof the ninth embodiment, wherein carrier intervals are not weighted;

FIG. 127 is a block diagram showing a transmitter/receiver according tothe ninth embodiment;

FIG. 128(a) is a block diagram of a trellis encoder (ratio ½) inembodiments 2, 4, and 5,

FIG. 128(b) is a block diagram of a trellis encoder (ratio ⅔) inembodiments 2, 4, and 5,

FIG. 128(c) is a block diagram of a trellis encoder (ratio ¾) inembodiments 2, 4, and 5,

FIG. 128(d) is a block diagram of a trellis decoder (ratio ½) inembodiments 2, 4, and 5,

FIG. 128(e) is a block diagram of a trellis decoder (ratio ⅔) inembodiments 2, 4, and 5,

FIG. 128(f) is a block diagram of a trellis decoder (ratio ¾) inembodiments 2, 4, and 5;

FIG. 129 is a view showing a time assignment of effective symbol periodsand guard intervals according to the ninth embodiment;

FIG. 130 is a graphic diagram showing a relationship between C/N rateand error rate according to the ninth embodiment;

FIG. 131 is a block diagram showing a magnetic recording/playbackapparatus according to the fifth embodiment;

FIG. 132 is a view showing a recording format of track on the magnetictape and a travellingtraveling of a head;

FIG. 133 is a block diagram showing a transmitter/receiver according tothe third embodiment;

FIG. 134 is a diagram showing a frequency assignment of a conventionalbroadcasting;

FIG. 135 is a diagram showing a relationship between service area andpicture quality in a three-level signal transmission system according tothe third embodiment;

FIG. 136 is a diagram showing a frequency assignment in case themulti-level signal transmission system according to the third embodimentis combined with FDM;

FIG. 137 is a block diagram showing a transmitter/receiver according tothe third embodiment, in which Trellis encoding is adopted;

FIG. 138 is a block diagram showing a transmitter/receiver according tothe ninth embodiment, in which a part of low frequency band signal istransmitted by OFDM;

FIG. 139 is a diagram showing an assignment of signal points of the8-PS-APSK signal of the first embodiment;

FIG. 140 is a diagram showing an assignment of signal points of the16-PS-APSK signal of the first embodiment;

FIG. 141 is a diagram showing an assignment of signal points of the8-PS-PSK signal of the first embodiment;

FIG. 142 is a diagram showing an assignment of signal points of the16-PS-PSK (PS type) signal of the first embodiment;

FIG. 143 is a graphic diagram showing the relationship between antennaradius of satellite and transmission capacity according to the firstembodiment;

FIG. 144 is a block diagram showing a weighted OFDM transmitter/receiveraccording to the ninth embodiment;

FIG. 145(a) is a diagram showing the waveform of the guard time and thesymbol time in the multi-level OFDM according to the ninth embodiment,wherein multipath is short;

FIG. 145(b) is a diagram showing the waveform of the guard time and thesymbol time in the multi-level OFDM according to the ninth embodiment,wherein multipath is long;

FIG. 146 is a diagram showing a principle of the multi-level OFDMaccording to the ninth embodiment;

FIG. 147 is a diagram showing subchannel assignment of a two-layersignal transmission system, weighted electric power according to theninth embodiment;

FIG. 148 is a diagram showing relationship among the D/V ratio, themultipath delay time, and the guard time according to the ninthembodiment;

FIG. 149(a) is a diagram showing time slots of respective layersaccording to the ninth embodiment;

FIG. 149(b) is a diagram showing time distribution of guard times ofrespective layers according to the ninth embodiment;

FIG. 149(c) is a diagram showing time distribution of guard times ofrespective layers according to the ninth embodiment;

FIG. 150 is a diagram showing the relationship between multipath delaytime and transfer rate according to the ninth embodiment, wherein athree-layer signal transmission effective to multipath is realized; and

FIG. 151 is a diagram showing the relationship between multipath delaytime and C/N ratio according to the ninth embodiment, whereintwo-dimensional, matrix type, multi-layer broadcast service can berealized by combining the GTW-OFDM and the C-CDM (or the CSW-OFDM).

FIG. 152 is a timing chart of a 3-level hierarchical television signalat each time slot when GTW-OFDM of the ninth embodiment is combined withC-CDM (or CSW-OFDM);

FIG. 153 shows the relationship between the multipath signal delay time,C/N ratio, and transmission rate when GTW-OFDM of the ninth embodimentis combined with C-CDM (or CSW-OFDM), and is used to describe thehierarchical broadcasting method using three-dimensional matrixstructure;

FIGS. 154A-C together form a frequency distribution graph of powerweight OFDM in the ninth embodiment;

FIG. 155 shows the position on the time axis of a 3-level hierarchicaltelevision signal at each time slot when guard time-OFDM of the ninthembodiment is combined with C-CDM;

FIG. 156 is a block diagram of the transmitter and the receiver in thefourth and fifth embodiments of the invention;

FIG. 157 is a block diagram of the transmitter and the receiver in thefourth and fifth embodiments of the invention;

FIG. 158 is a block diagram of the transmitter and the receiver in thefourth and fifth embodiments of the invention;

FIG. 159(a) is a signal point positioning diagram in 16-level VSB in thefifth embodiment of the invention;

FIG. 159(b) is a signal point positioning (8-level VSB) diagram in16-level VSB in the fifth embodiment of the invention;

FIG. 159(c) is a signal point positioning (4-level VSB) diagram in16-level VSB in the fifth embodiment of the invention;

FIG. 159(d) is a signal point positioning (16-level VSB) diagram in16-level VSB in the fifth embodiment of the invention;

FIG. 160(a) is a block diagram of an ECC encoder in the fifth and sixthembodiments of the invention;

FIG. 160(b) is a block diagram of an ECC decoder in the fifth and sixthembodiments of the invention;

FIG. 161 is an overall block diagram of a VSB receiver in the fifthembodiment of the invention;

FIG. 162 is a block diagram of a the receiver in the fifth embodiment ofthe invention;

FIG. 163 is a graph of the error rate and C/N ratio curve in 4-level VSBand TC-8-level VSB in the fourth embodiment of the invention;

FIG. 164 is an error rate curve of subchannel 1 and subchannel 2 in4-level VSB and TC-8-level VSB in the fourth embodiment of theinvention;

FIG. 165(a) is a block diagram of the Reed-Solomon encoder in thesecond, fourth, and fifth embodiments of the invention;

FIG. 165(b) is a block diagram of the Reed-Solomon decoder in thesecond, fourth, and fifth embodiments of the invention;

FIG. 166 is a flowchart of Reed-Solomon error correction and operationin the second, fourth and fifth embodiments of the invention;

FIG. 167 is a block diagram of the deinterleaver in the second, third,fourth, fifth and sixth embodiments of the invention;

FIG. 168(a) is an interleave/deinterleave table for the second, third,fourth, and fifth embodiments of the invention;

FIG. 168(b) shows the interleave distance in the second, third, fourth,and fifth embodiments of the invention;

FIG. 169 is a comparison of redundancy in 4-level VSB, 8-level VSB, and16-level VSB in the fifth embodiment of the invention;

FIG. 170 is a block diagram of a television receiver for receiving thehigh priority signal of the second, third, fourth, and fifth embodimentsof the invention;

FIG. 171 is a block diagram of the receiver and transmitter in thesecond, third, fourth, and fifth embodiments of the invention;

FIG. 172 is a block diagram of the receiver and transmitter in thesecond, third, fourth, and fifth embodiments of the invention; and

FIG. 173 is a block diagram of an ASK magnetic recording and reproducingapparatus according to the sixth embodiment of the invention;

FIG. 174 is a block diagram showing a circuitry arrangement of QAM/VSBcompatible modulator for multi-level transmission according toEmbodiment 5.

FIG. 175 is a block diagram showing another circuitry arrangement of theQAM/VSB modulator for multi-level transmission according to Embodiment5.

FIG. 176 illustrates a third modification of the QAM/VSB modulator ofEmbodiment 5.

FIG. 177 is a block diagram showing a Trellis decoder in the demodulatorof Embodiment 5.

FIG. 178 is a block diagram of a receiver of Embodiment 5 forinterception of VSB multi-level transmitted signals emitted in the air.

FIG. 179is illustrates another arrangement of the QAM/VSB compatiblereceiver of Embodiment 5.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS EMBODIMENT 1

One embodiment of the present invention will be described referring tothe relevant drawings.

In the preferred embodiment of the invention both the transmissionapparatus, which comprises a transmitter for transmitting a digital HDTVsignal or other digital signal and a receiver for receiving thetransmitted signal, and the recording and reproducing apparatus, whichrecords the digital HDTV signal or other digital signal on a magnetictape or other recording medium and reproduces the recorded signal fromsaid medium, are described.

It should be noted, however, that the configuration, operation, andprinciple of the digital modulator and demodulator, error correctionencoder and decoder, and the encoder and decoder for image coding theHDTV signal are common to the transmission apparatus and the recordingand reproducing apparatus, and apply essentially the same technologies.Therefore, to more concisely describe each embodiment, the blockdiagrams for either the transmission apparatus or the recording andreproducing apparatus are referenced in the description of eachembodiment. In addition, the configuration of each embodiment of theinvention can be achieved by means of any multi-value digital modulationmethod, e.g., QAM, ASK and PSK, positioning signal points in aconstellation, and for brevity the embodiments of the present inventionare described using only one modulation method. FIG. 1 shows the entirearrangement of a signal transmission system according to the firstembodiment of the present invention. A transmitter 1 comprises an inputunit 2, divider circuit 3, a modulator 4, and a transmitter unit 5. Inoperation, each input multiplex signal is divided by the divider circuit3 into three groups, a first data stream D1, a second data stream and, athird data stream D3, which are then modulated by the modulator 4 beforebeing transmitted from the transmitter unit 5. The modulated signal issent up from an antenna 6 through an uplink 7 to a satellite 10 where itis intercepted by an uplink antenna 11 and amplified by a transponder 12before being transmitted from a downlink antenna 13 towards the ground.

The transmission signal is then sent down through three downlinks 21, 3231, and 41 to a first 23, a second 33, and a third receiver 43respectively. In the first receiver 23, the signal intercepted by anantenna 22 is fed through an input unit 24 to a demodulator 25 where itsfirst data stream only is demodulated, while the second and third datastreams are not recovered, before being transmitted further from anoutput unit 26.

Similarly, the second receiver 33 allows the first and second datastreams of the signal intercepted by an antenna 32 and fed from an inputunit 34 to be demodulated by a demodulator 35 and then, combined by amixer 37 into a single data stream which is then transmitted furtherfrom an output unit 36.

The third receiver 43 allows all of the first, second, and third datastreams of the signal intercepted by an antenna 42 and fed from an inputunit 44 to be demodulated by a demodulator 45 and then, combined by amixer 47 into a single data stream which is then transmitted furtherfrom an output unit 46.

As understood, the three discrete receivers 23, 33, and 43 have theirrespective demodulators of different characteristics such that theiroutputs demodulated from the same frequency band signal of thetransmitter 1 contain data of different sizes. More particularly, threedifferent but compatible data can simultaneously be carried on a givenfrequency band signal to their respective receivers. For example, eachof three, existing NTSC, HDTV, and super HDTV, digital signals isdivided into low, high, and super high frequency band components whichrepresent the first, the second, and the third data stream respectively.Accordingly, the three different TV signals can be transmitted on aone-channel frequency band carrier for simultaneous reproduction ofmedium, high, and super high resolution TV images respectively.

The NTSC TV signal is intercepted by a receiver accompanied by a smallantenna for demodulation of small-sized data; the HDTV signal isintercepted by a receiver accompanied by a medium antenna fordemodulation of medium-sized data, and the super HDTV signal isintercepted by a receiver accompanied by a large antenna fordemodulation of large-sized data. Also, as illustrated in FIG. 1, adigital NTSC TV signal containing only the first data stream for digitalNTSC TV broadcasting service is fed to a digital transmitter 51 where itis received by an input unit 52 and modulated by a demodulator modulator54 before being transmitted further from a transmitter unit 55. Thedemodulated modulated signal is then sent up from an antenna 56 throughan uplink 57 to the satellite 10 which in turn transmits the samethrough a downlink 58 to the first receiver 23 on the ground.

The first receiver 23 demodulates with its demodulator 25 the modulateddigital signal supplied from the digital transmitter 51 into theoriginal first data stream signal. Similarly, the same modulated digitalsignal can be intercepted and demodulated by the second receiver 33 orthird receiver 43 into the first data stream or NTSC TV signal. Insummary, the three discrete receivers 23, 33, and 43 all can interceptand process a digital signal of the existing TV system for reproduction.

The arrangement of the signal transmission system will be described inmore detail.

FIG. 2 is a block diagram of the transmitter 1, in which an input signalis fed across the input unit 2 and divided by the divider circuit 3 intothree digital signals containing a first, a second, and a third datastream respectively.

Assuming that the input signal is a video signal, its low frequency bandcomponent is assigned to the first data stream, its high frequency bandcomponent to the second data stream, its super-high frequency bandcomponent to the third data stream. The three different frequency bandsignals are fed to a modulator input 61 of the modulator 4. Here, asignal point shifting circuit 67 shifts the positions of the signalpoints according to an externally given signal. The modulator 4 isarranged for amplitude modulation on two 90°-out-of phase carriersrespectively which are then combined into a multiple QAM signal. Morespecifically, the signal from the modulator input 61 is fed to both afirst AM modulator 64 62 and a second AM modulator 63. Also, a carrierwave of cos(2πfct) produced by a carrier generator 64 is directly fed tothe first AM modulator 64 62 and also, to a π/2 phase shifter 66 whereit is 90° shifted in phase to a sin(2πfct) form prior to beingtransmitted to the second AM modulator 63. The two amplitude modulatedsignals from the first and second AM modulators 64 62, 63 are combinedby a summer 65 into a transmission signal which is then transferred tothe transmitter unit 5 for output. The procedure is well known and willnot be further explained.

The QAM signal will now be described in a common 4×4 or 16 stateconstellation referring to the first quadrant of a space diagram in FIG.3. The output signal of the modulator 4 is expressed by a sum vector oftwo, Acos2πfct and Bcos Bsin 2πfct, vectors 81 and 82 which respectivelyrepresent the two 90°-out-of-phase carriers. When the distal point of asum vector from the zero point represents a signal point, the 16 QAMsignal has 16 signal points determined by a combination of fourhorizontal amplitude values a₁, a₂, a₃, and a₄ and four verticalamplitude values b₁, b₂, b₃, and b₄. The first quadrant in FIG. 3contains four signal points 83 at c₁₁, 84 at c₁₂, 85 at c₂₂, and 86 atc₂₁.

c₁₁ is a sum vector of a vector 0-a₁ and a vector 0-b₁ and thus,expressed as c₁₁=a₁ cos2πfct-b₁ sin2πfct=Acos (2πfct+dπ/2).

It is now assumed that the distance between 0 and a₁ in the orthogonalcoordinates of FIG. 3 is A₁, between a₁ and a₂ is A₂, between 0 and b₁is B₁, and between b₁ and b₂ is B₂.

As shown in FIG. 4, the 16 signal points are allocated in a vectorcoordinate, in which each point represents a four-bit pattern thus toallow the transmission of four bit data per period or time slot.

FIG. 5 illustrates a common assignment of two-bit patterns to the 16signal points.

When the distance between two adjacent signal points is great, it willbe identified by the receiver with much ease. Hence, it is desirable tospace the signal points at greater intervals. If two particular signalpoints are allocated near to each other, they are rarely distinguishedand the error rate will be increased. Therefore, it is most preferableto have the signal points spaced at equal intervals as shown in FIG. 5,in which the 16 QAM signal is defined by A₁=A₂/2.

The transmitter 1 of the embodiment is arranged to divide an inputdigital signal into a first, a second, and a third data or bit stream.The 16 signal points or groups of signal points are divided into fourgroups. Then, 4 two-bit patterns of the first data stream are assignedto the four signal point groups respectively, as shown in FIG. 6. Moreparticularly, when the two-bit pattern of the first data stream is 11,one of four signal points of the first signal point group 91 in thefirst quadrant is selected depending on the content of the second datastream for transmission. Similarly, when 01, one signal point of thesecond signal point group 92 in the second quadrant is selected andtransmitted. When 00, one signal point of the third signal point group93 in the third quadrant is transmitted and when 10, one signal point ofthe fourth signal point group 94 in the fourth quadrant is transmitted.Also, 4 two-bit patterns in the second data stream of the 16 QAM signal,or e.g. 16 four-bit patterns in the second data stream of a 64-state QAMsignal, are assigned to four signal points or sub signal point groups ofeach of the four signal point groups 91, 92, 93, and 94 respectively, asshown in FIG. 7. It should be understood that the assignment issymmetrical between any two quadrants. The assignment of the signalpoints to the four groups 91, 92, 93, and 94 is determined by priorityto the two-bit data of the first data stream. As the result, two-bitdata of the first data stream and two-bit data of the second data streamcan be transmitted independently. Also, the first data stream will bedemodulated by using a common 4 PSK receiver having a given antennasensitivity. If the antenna sensitivity is higher, a modified type ofthe 16 QAM receiver of the present invention will intercept anddemodulate both the first and second data stream streams with equalsuccess.

FIG. 8 shows an example of the assignment of the first and second datastreams in two-bit patterns.

When the low frequency band component of an HDTV video signal isassigned to the first data stream and the high frequency component tothe second data stream, the 4 PSK receiver can produce an NTSC-levelpicture from the first data stream and the 16- or 64-state QAM receivercan produce an HDTV picture from a composite reproduction signal of thefirst and second data streams.

Since the signal points are allocated at equal intervals, there isdeveloped in the 4 PSK receiver a threshold distance between thecoordinate axes and the shaded area of the first quadrant, as shown inFIG. 9. If the threshold distance is A_(T0), a PSK signal having anamplitude of A_(T0) will successfully be intercepted. However, theamplitude has to be increased to a three times greater value of 3A_(T0)for transmission of a 16 QAM signal while the threshold distance A_(T0)is maintained. More particularly, the energy needed for transmitting the16 QAM signal is nine times greater than that for sending the 4 PSKsignal. Also, when the 4 PSK signal is transmitted in a 16 QAM mode,energy waste will be high and reproduction of a carrier signal will betroublesome. Above all, the energy available for satellite transmittingis not abundant but strictly limited to minimum use. Hence, nolarge-energy-consuming signal transmitting system will be put intopractice until more energy for satellite transmission is available. Itis expected that a great number of the 4 PSK receivers will beintroduced into the market as digital TV broadcasting is placed inservice. After introduction to the market, the 4 PSK receivers willhardly be shifted to higher sensitivity models because a signalintercepting characteristic gap between the two, old and new, models ishigh. Therefore, the transmission of the 4 PSK signals must not beabandoned. In this respect, a new system is desperately needed fortransmitting the signal point data of a quasi 4 PSK signal in the 16 QAMmode using less energy. Otherwise, the limited energy at a satellitestation will degrade the entire transmission system.

The present invention resides in a multiple signal level arrangement inwhich the four signal point groups 91, 92, 93, and 94 are allocated at agreater distance from each other, as shown in FIG. 10, for minimizingthe energy consumption required for 16 QAM modulation of quasi 4 PSKsignals.

For clearing the relationship between the signal receiving sensitivityand the transmitting energy, the arrangement of the digital transmitter51 and the first receiver 23 will be described in more detail referringto FIG. 1.

Both the digital transmitter 51 and the first receiver 2 3 23 are formedof known types for data transmission or video signal transmission e.g.in TV broadcasting service. As shown in FIG. 17, the digital transmitter51 is a 4 PSK transmitter equivalent to the multiple-bit QAM transmitter1, shown in FIG. 2, without AM modulation capability. In operation, aninput signal is fed through an input unit 52 to a modulator 54 where itis divided by a modulator input 121 into two components. The twocomponents are then transferred to a first two-phase modulator circuit122 for phase modulation of a base carrier and a second two-phasemodulator circuit 123 for phase modulation of a carrier which is 90° outof phase with the base carrier respectively. The two outputs of thefirst and second two-phase modulator circuits 122 and 123 are thencombined by a summer 65 into a composite modulated signal which isfurther transmitted from a transmitter unit 55.

The resultant modulated signal is shown in the space diagram of FIG. 18.

It is known that the four signal points are allcated allocated at equaldistances for achieving optimum energy utilization. FIG. 18 illustratesan example where the four signal points 125, 126, 127, and 128 represent4 two-bit patterns, 11, 01, 00, and 10 respectively. It is alsodesirable for successful data transfer from the digital transmitter 51to the first receiver 23 that the 4 PSK signal from the digitaltransmitter 51 has an amplitude of not less than a given level. Morespecifically, when the minimum amplitude of the 4 PSK signal needed fortransmission from the digital transmitter 51 to the first receiver 23 of4 PSK mode, or the distance between 0 and a₁ in FIG. 18 is A_(T0), thefirst receiver 23 must successfully intercept any 4 PSK signal having anamplitude of more than A_(T0).

The first receiver 23 is arranged to receive at its small-diameterantenna 22 a desired or 4 PSK signal which is transmitted from thetransmitter 1 or digital transmitter 51 respectively through thetransponder 12 of the satellite 10 and demodulate it with thedemodulator 24 25. In more detail, the first receiver 23 issubstantially designed for interception of a digital TV or datacommunications signal of 4 PSK or 2 PSK mode.

FIG. 19 is a block diagram of the first receiver 23 in which an inputsignal received by the antenna 22 from the satellite 12 10is fed throughthe input unit 24 to a carrier reproducing circuit 131 where a carrierwave is demodulated and to a π/2 phase shifter 132 where a 90° phasecarrier wave is demodulated. Also, two 90°-out-of-phase components ofthe input signal are respectively detected by a first phase detectorcircuit 133 and a second phase detector circuit 134 and are respectivelytransferred to first 136 and second discrimination/demodulation circuits136 and 137. Two demodulated components from their respectivediscrimination/demodulation circuits 136 and 137, which have separatelybeen discriminated at units of time slot by means of timing signals froma timing wave extracting circuit 135, are fed to a first data streamreproducing unit 232 where they are combined into a first data streamsignal which is then delivered as an output from the output unit 26.

The input signal to the first receiver 23 will now be explained in moredetail referring to the vector diagram of FIG. 20. The 4 PSK signalreceived by the first receiver 23 from the digital transmitter 51 isexpressed in an ideal form without transmission distortion and noise,using four signal points 151, 152, 153, and 154, as shown in FIG. 20.

In practice, the real four signal points appear in particular extendedareas about the ideal signal positions 151, 152, 153, and 154respectively due to noise, amplitude distortion, and phase errordeveloped during transmission. If one signal point is unfavorablydisplaced from its original position, it will hardly be distinguishedfrom its neighboring signal point and the error rate will thus beincreased. As the error rate increases to a critical level, thereproduction of data becomes less accurate. For enabling the datareproduction at a maximum acceptable level of the error rate, thedistance between any two signal points should be far enough to bedistinguished from each other. If the distance is 1A_(R0), the signalpoint 151 of a 4 PSK signal close to a critical error level has to stayin a first discrimination area 155 denoted by the hatching of FIG. 20and determined by |0-a_(R1)|>A_(R0) and |0-b_(R1)|>A_(R0). This allowsthe signal transmission system to reproduce carrier waves and thus,demodulate a wanted signal. When the minimum radius of the antenna 22 isset to r₀, the transmission signal of more than a given level can beintercepted by any receiver of the system. The amplitude of a 4 PSKsignal of the digital transmitter 51 shown in FIG. 18 is minimum atA_(T0) and thus, the minimum amplitude A_(R0) of a 4 PSK signal to bereceived by the first receiver 23 is determined to be equal to A_(T0).As a result, the first receiver 23 can intercept and demodulate the 4PSK signal from the digital transmitter 41 at the maximum acceptablelevel of the error rate when the radius of the antenna 22 is more thanr₀. If the transmission signal is of a modified 16- or 64-state QAMmode, the first receiver 23 may find it difficult to reproduce itscarrier wave. For compensation, the signal points are increased to eightwhich are allocated at angles of (π/4+nπ/2) as shown in FIG. 25(a) andits carrier wave will be reproduced by a 16× multiplication technique.Also, if the signal points are assigned to 16 locations at angles ofnπ/8 as shown in FIG. 25(b), the carrier of a quasi 4 PSK mode 16 QAMmodulated signal can be reproduced with the carrier reproducing circuit131 which is modified for performing 16× frequency multiplication. Atthe time, the signal points in the transmitter 1 should be arranged tosatisfy A₁/(A₁+A₂)=tan(π/8).

Here, a case of receiving a QPSK signal will be considered. Similarly tothe manner performed by the signal point setting circuit 67 in thetransmitter shown in FIG. 2, it is also possible to modulate thepositions of the signal points of the QPSK signal shown in FIG. 18(amplitude-modulation, pulse-modulation, or the like). In this case, thesignal point demodulating unit 138 in the first receiver 23 demodulatesthe position modulated or position changed signal. The demodulatedsignal is outputted together with the first data stream.

The 16 PSK signal of the transmitter 1 will now be explained referringto the vector diagram of FIG. 9. When the horizontal vector distance A₁of the signal point 83 is greater than A_(T0) of the minimum amplitudeof the 4 PSK signal of the digital transmitter 51, the four signalpoints 83, 84, 85, and 86 in the first quadrant of FIG. 9 stay in theshaded or first 4 PSK signal receivable area 87. When received by thefirst receiver 23, the four points of the signal appear in the firstdiscriminating area of the vector field shown in FIG. 20. Hence, any ofthe signal points 83, 84, 85, and 86 of FIG. 9 can be translated intothe signal level 151 of FIG. 20 by the first receiver 23 so that thetwo-bit pattern of 11 is assigned to a corresponding time slot. Thetwo-bit pattern of 11 is identical to 11 of the first signal point group91 or first data stream of a signal from the transmitter 1. Equally, thefirst data stream will be reproduced at the second, third, or fourthquadrant. As the result, the first receiver 23 reproduces two-bit dataof the first data stream out of the plurality of data streams in a 16-,32-, or 64-state QAM signal transmitted from the transmitter 1. Thesecond and third data streams are contained in four segments of thesignal point group 91 and thus, will not affect the demodulation of thefirst data stream. They may however affect the reproduction of a carrierwave and an adjustment, described later, will be needed.

If the transponder of a satellite supplies an abundance of energy, theforgoing technique of 16 to 64-state QAM mode transmission will befeasible. However, the transponder of the satellite in any existingsatellite transmission system is strictly limited in the power supplydue to its compact size and the capability of solar batteries. If thetransponder or satellite is increased in size and thus weight, itslaunching cost will soar. This disadvantage will rarely be eliminated bytraditional techniques unless the cost of launching a satellite rocketis reduced by to a considerable level. In the existing system, a commoncommunications satellite provides as low as 20 W of power and a commonbroadcast satellite offers 100 W to 200 W at best. For transmission ofsuch a 4 PSK signal in the symmetrical 16-state QAM mode as shown inFIG. 9, the minimum signal point distance is needed needed is 3A_(T0) asthe 16 QAM amplitude is expressed by 2A₁=A₂. Thus, the energy needed forthe purpose is nine times greater than that for transmission of a common4 PSK signal, in order to maintain compatibility. Also, any conventionalsatellite transponder can hardly provide a power for enabling such asmall antenna of the 4 PSK first receiver to intercept a transmittedsignal therefrom. For example, in the existing 40 W system, 360 W isneeded for appropriate signal transmission and will be unrealistic withrespect to cost.

It would be understood that the symmetrical signal state QAM techniqueis most effective when the receivers equipped with the same sizedantennas are employed corresponding to a given transmitting power.Another novel technique will however be preferred for use with receiversequipped with different sized antennas.

In more detail, while the 4 PSK signal can be intercepted by a commonlow cost receiver system having a small antenna, the 16 QAM signal isintended to be received by a high cost, high quality, multiple-bitmodulating receiver system with a medium or large sized antenna which isdesigned for providing highly valuable services, e.g. HDTVentertainment, to a particular person who invests more money. Thisallows both 4 PSK and 16 QAM signals, if desired, with a 64 DMA QAM, tobe transmitted simultaneously with the help of a small increase in thetransmitting power.

For example, the transmitting power can be maintained low when thesignal points are allocated at A₁=A₂ as shown in FIG. 10. The amplitudeA(4) for transmission of 4 PSK data is expressed by a vector 96equivalent to the square root of (A₁+A₂)²+(B₁+B₂)². Then,|A(4)|₂=A₁ ²+B₁ ²=A² _(T0)+A² _(T0)=2A² _(T0)|A(16)|₂=(A₁+A₂)²+(B₁+B₂)²=4A² _(T0)+4A² _(T0)=8A_(T0)|A(16)|/|A(4)|=2

Accordingly, the 16 QAM signal can be transmitted at a two times greateramplitude and a four times greater transmitting energy than those neededfor the 4 PSK signal. A modified 16 QAM signal according to the presentinvention will not be demodulated by a common receiver designed forsymmetrical, equally distanced signal point QAM. However, it can bedemodulated with the second receiver 33 when two threshold values A₁ andA₂ are preset to appropriate values. In FIG. 10, the minimum distancebetween two signal points in the first segment of the signal point group91 is A₁ and A₂/2A₁ is established as compared with the distance 2A₁ of4 PSK. Then, as A₁=A₂, the distance becomes ½. This explains that thesignal receiving sensitivity has to be two times greater for the sameerror rate and four times greater for the same signal level. For havinga four times greater value of sensitivity, the radius r₂ of the antenna32 of the second receiver 33 has to be two times greater than the radiusr₁ of the antenna 22 of the first receiver 23 thus satisfying r₂=2r₁.For example, the antenna 32 of the second receiver 33 is 60 cm diameterwhen the antenna 22 if the first receiver 23 is 30 cm. In this manner,the second data stream representing the high frequency component of anHDTV will be carried on a signal channel and demodulated successfully.As the second receiver 33 intercepts the second data stream or a higherdata signal, its owner can enjoy a of high return of investment return .Hence, the second receiver 33 of a high price may be accepted. As theminimum energy for transmission of 4 PSK data is predetermined, theratio n₁₆ of modified 16 APSK transmitting energy to 4 PSK transmittingenergy will be calculated according to the antenna radius r₂ of thesecond receiver 33 using a ratio between A₁ and A₂ shown in FIG. 10.

In particular, n₁₆ is expressed by ((A₁+A₂)/A₁)² which is the minimumenergy for transmission of 4 PSK data. As the signal point distancesuited for modified 16 QAM interception is A₂, The the signal pointdistance for 4 PSK interception is 2A₁, and the signal point distanceratio is A₂/2A₁, the antenna radius r₂ is determined as shown in FIG.11, in which the curve 101 represents the relationship between thetransmitting energy ratio n₁₆ and the radius r₂ of the antenna 22 of thesecond receiver 23.

Also, the point 102 indicates transmission of common 16 QAM at the equaldistance signal state mode where the transmitting energy is nine timesgreater and thus will no more be practical. As apparent from the graphof FIG. 11, the antenna radius r₂ of the second receiver 23 cannot bereduced further even if n₁₆ is increased more than 5 times.

The transmitting energy at the satellite is limited to a small value andthus, n₁₆ preferably stays not more than 5 times the value, as denotedby the hatching of FIG. 11. The point 104 within the hatching area 103indicates, for example, that the antenna radius r₂ of a two timesgreater value is matched with a 4× value of the transmitting energy.Also, the point 105 represents that the transmission energy should bedoubled when r₂ is about 5× greater. Those values are all within afeasible range.

The value of n₁₆ not greater than 5× value is expressed using A₁ and A₂as:n₁₆=((A₁+A₂)/A₁)²<5Hence, A₂<1.23A₁.

If the distance between any two signal point group segments shown inFIG. 10 is 2A(4) and the maximum amplitude is 2A(16), A(4) andA(16)-A(4) are proportional to A1 A₁ and A2 A₂ respectively. Hence,(A(16))²<5(A(14))² is established.

The action of a modified 64 ASPK transmission will be described as thethird receiver 43 can perform 64-state QAM demodulation.

FIG. 12 is a vector diagram in which each signal point group segmentcontains 16 signal points as compared with 4 signal points of FIG. 10.The first signal point group segment 91 in FIG. 12 has a 4×4 matrix of16 signal points allocated at equal intervals including the point 170.For providing compatibility with 4 PSK, A₁>A_(T0) has to be satisfied.If the radius of the antenna 42 of the third receiver 43 is r₃ and thetransmitting energy is n₆₄, the equation is expressed as:r₃ ²={6²/(n−1)}r₁ ²

This relationship between r³ r₃ and n of a 64 QAM signal is also shownin the graphic representation of FIG. 13.

It is understood that the signal point assignment shown in FIG. 12allows the second receiver 33 to demodulate only two-bit patterns of 4PSK data. Hence, it is desirable for to have compatibility between amongthe first, second, and third receivers that the second receiver 33 iscapable of demodulating a modified 16 QAM form from the 64 QAM modulatedsignal.

The compatibility between among the three discrete receivers can beimplemented by a three-level grouping of signal points, as illustratedin FIG. 14. A description follows referring to the first quadrant inwhich the first signal point group segment 91 represents the two-bitpattern 11 of the first data stream.

In particular, a first sub segment 181 in the first signal point groupsegment 91 is assigned the two-bit pattern 11 of the second data stream.Equally, a second 182, a third 183, and a fourth sub segment 184 areassigned 01, 00, and 10 of the same respectively. This assignment isidentical to that shown in FIG. 7.

The signal point allocation of the third data stream will now beexplained referring to the vector diagram of FIG. 15 which shows thefirst quadrant. As shown, the four signal points 201, 205, 209, and 213represent the two-bit pattern of 11, the signal points 202, 206, 210,and 214 represent 01, the signal points 203, 207, 211, and 215 represent00, and signal points 204, 208, 212, and 216 represent 10. Accordingly,the two-bit patterns of the third data stream can be transmittedseparately of the first and second data streams. In other words, two-bitdata of the three different signal levels can be transmittedrespectively.

As understood, the present invention permits not only transmission ofsix-bit data but also interception of three, two-bit, four-bit, andsix-bit, different bit length data with their respective receivers whilethe signal compatibility remains between these levels.

The signal point allocation for providing compatibility between amongthe three levels will be described.

As shown in FIG. 15, A₁>A_(T0) is essential for allowing the firstreceiver 23 to receive the first data stream.

It is necessary to space any two signal points from each other by such adistance that the sub segment signal points, e.g. 182, 183, 184, of thesecond data stream shown in FIG. 15 can be distinguished from the signalpoint 91 shown in FIG. 10.

FIG. 15 shows that they are spaced by ⅔A₂. In this case, the distancebetween the two signal points 201 and 202 in the first sub segment 181is A₂/6. The transmitting energy needed for signal interception with thethird receiver 43 is now calculated. If the radius of the antenna 3242is r₃ and the needed transmitting energy is n₆₄ times the 4 PSKtransmitting energy, the equation is expressed as:R₃ ²(12r₁)²/(n−1)R₃ ² =(12r ₁ ) ² /(n−1)

This relationship is also denoted by the curve 211 in FIG. 16. Forexample, if the transmitting energy is 6 or 9 times greater than thatfor 4 PSK transmission at the point 223 or 222, the antenna 32 having aradius of 8× or 6× value respectively can intercept the first, second,and third data streams for demodulation. As the signal point distance ofthe second data stream is close to ⅔A₂, the relationship between r₁ andr₂ is expressed by:R₂ 2=(3r₁)²/(n−1) R₂ ² =(3r ₁ ) ² /(n−1 )

Therefore, the antenna 32 of the second receiver 33 has to be slightlyincreased in radius as denoted by the curve 223.

As understood, while the first and second data streams are transmittedthrough a traditional satellite which provides a small signaltransmitting energy, the third data stream can also be transmittedthrough a future satellite which provides a greater signal transmittingenergy without interrupting the action of the first and second receivers23 or 33 or with no need of modification of the same and thus, both thecompatibility and the advancement is ensured.

The signal receiving action of the second receiver 33 will first bedescribed. As compared with the first receiver 23 arranged forinterception with a small radius r₁ antenna and demodulation of the 4PSK modulated signal of the digital transmitter 51 or the first datastream of the signal of the transmitter 1, the second receiver 33 isadopted for perfectly demodulating the 16 signal state two-bit data,shown in FIG. 10, or second data stream of the 16 QAM signal from thetransmitter 1. In total, four-bit data including also the first datastream can be demodulated. The ratio between A₁ and A₂ is howeverdifferent in the two transmitters. The two different data are loaded toa demodulation controller 231 of the second receiver 33, shown in FIG.21, which in turn supplies their respective threshold values to thedemodulating circuit for AM demodulation.

The block diagram of the second receiver 33 in FIG. 21 is similar inbasic construction to that of the first receiver 23 shown in FIG. 19.The difference is that the radius r₂ of the antenna 32 is greater thanr₁ of the antenna 22. This allows the second receiver 33 to identify asignal component involving a smaller signal point distance. Thedemodulator 35 of the second receiver 33 also contains first and seconddata stream reproducing units 232 and 233 in addition to thedemodulation controller 231. There is provided a firstdiscrimination/demodulation circuit 136 for AM demodulation of modified16 QAM signals. As understood, each carrier is a four-bit signal havingtwo, positive and negative, threshold values about the zero level. AS Asapparent from the vector diagram, of FIG. 22, the threshold values arevaried depending on the transmitting energy of a transmitter since thetransmitting signal of the embodiment is a modified 16 QAM signal. Whenthe reference threshold is TH₁₆, it is determined by, as shown in FIG.22:TH₁₆=(A₁+A₂/2)/(A₁+A₂)

The various data for demodulation including A₁ and A₂ or TH₁₆, and thevalue m for multiple-bit modulation are also transmitted from thetransmitter 1 as carried in the first data stream. The demodulationcontroller 231 may be arranged for recovering such demodulation datathrough statistical process of the received signal.

A way of determining the shift factor A₁/A₂ will be described withreference to FIG. 26. A change of the shift factor A₁/A₂ causes a changeof the threshold value. Increase of a difference of a value of A₁/A₂ setat the receiver side from a value of A₁/A₂ set at the transmitter sidewill increase the error rate. Referring to FIG. 26, the demodulatedsignal from the second data stream reproducing unit 233 may be fed backto the demodulation controller 231 to change the shift factor A₁/A₂ in adirection to increase the error rate. By this arrangement, the thirdreceiver 43 may not demodulate the shift factor A₁/A₂, so that thecircuit construction can be simplified. Further, the transmitter may nottransmit the shift factor A₁/A₂, so that the transmission capacity canbe increased. This technique can be applied also to the second receiver33.

FIGS. 25(a) and 25(b) are views showing signal point allocations for theC-CDM signal points, wherein signal points are added by shifting in thepolar coordinate direction (r, θ). The previously described C-CDM ischaracterized in that the signal points are shifted in the rectangularcoordinate direction, i.e. XY direction; therefore it is referred to asrectangular coordinate system C-CDM. Meanwhile, this C-CDM characterizedby the shifting of signal points in the polar coordinate direction, i.e.r, θ direction, is referred to as polar coordinate system C-CDM.

FIG. 25(a) shows the signal allocation of 8PS-APSK signals, wherein foursignal points are added by shifting each of 4 QPSK signals in the radiusr direction of the polar coordinate system. In this manner, the APSK ofpolar coordinate system C-CDM having 8 signal points is obtained fromthe QPSK as shown in FIG. 25(a). As the pole is shifted in the polarcoordinate system to add signal points in this APSK, it is referred toas shifted pole-APSK, i.e. SP-APSK in the abbreviated form. In thiscase, coordinate values of the newly added four QPSK signals 85 arespecified by using a shift factor S₁ as shown in FIG. 139. Namely,8PS-APSK signal points includes ordinary QPSK signal points 83 (r₀, θ₀)and a signal points ((S₁+1)(r₀, θ₀) obtained by shifting the signalpoint 83 in the radius r direction by an amount of S₁r₀. Thus, a 1-bitsubchannel 2 is obtained in addition to a 2-bit subchannel 1 identicalwith the QPSK.

Furthermore, as shown in the constellation diagram of FIG. 140, neweight signal points, represented by coordinates (r₀+S₂r₀, θ₀) and(r₀+S₁r₀+S₂r₀, θ₀), can be added by shifting the eight signal points(r₀, θ₀) and (r₀+S₁r₀, θ₀) in the radius r direction. As this allows twokinds of allocations, a 1-bit subchannel is obtained and is referred toas 16PS-APSK which provides the 2-bit subchannel 1, 1-bit subchannel 2,and 1-bit subchannel 3. As the 16-PS-APSK disposes the signal points onthe lines of θ=¼(2n+1)π, it allows the ordinary QPSK receiver explainedwith reference to FIG. 19 to reproduce the carrier wave to demodulatethe first 2-bit subchannel although the second subchannel cannot bedemodulated. As described above, the C-CDM method of shifting the signalpoints in the polar coordinate direction is useful in expanding thecapacity of information data transmission while assuring compatibilityto the PSK, especially to the QPSK receiver, a main receiver for thepresent satellite broadcast service. Therefore, without losing the firstgeneration viewers of the satellite broadcast service based on the PSK,the broadcast service will advance to a second generation stage whereinthe APSK will be used to increase transmittable information amount byuse of the multi-level modulation while maintaining compatibility.

In FIG. 25(b), the signal points are allocated on the lines of θ=π/8.With this arrangement, the 16 PSK signal points are reduced or limitedto 12 signal points, i.e. 3 signal points in each quadrant. With thislimitation, these three signal points in each quadrant are roughlyregarded as one signal point for 4 QPSK signals. Therefore, this enablesthe QPSK receiver to reproduce the first subchannel in the same manneras in the previous embodiment.

More specifically, the signal points are disposed on the lines of θ=π/4,θ=π/4+π/8, and θ=π/4−π/8. In other words, the added signals are offsetby an amount of ±θ in the angular direction of the polar coordinatesystem from the QPSK signals disposed on the lines of θ=π/4. Since allthe signals are in the range of θ=π/4±π/8, they can be regarded as oneof the QPSK signal points on the line of θ=π/4. Although the error rateis lowered a little bit in this case, the QPSK receiver 23 shown in FIG.19 can discriminate these points as four signal points angularlyallocated. Thus, 2-bit data can be reproduced. In case of the angularshift C-CDM, if signal points are disposed on the lines of π/n, thecarrier wave reproduction circuit can reproduce the carrier wave by theuse of an n-multiplier circuit in the same manner as in otherembodiments. If the signal points are not disposed on the lines of π/n,the carrier wave can be reproduced by transmitting several pieces ofcarrier information within a predetermined period in the same manner asin other embodiment embodiments. Assuming that an angle between twosignal points of the QPSK or 8-SP-APSK is 2θ₀ in the polar coordinatesystem and a first angular shift factor is P1, two signal points (r₀,θ₀+P₁θ₀) and (r₀, % θ₀-P₁θ₀) are obtained by shifting the QPSK signalpoint in the angular θ direction by an amount ± +P₁θ₀. Thus, the numberof signal points are doubled. Thus, the 1-bit subchannel 3 can be addedand is referred to as 8-SP-PSK of P=P1. If eight signal points arefurther added by shifting the 8-SP-PSK signals in the radius r directionby an amount S₁r₀, it will become possible to obtain 16-SP-APSK (P, S₁type) as shown in FIG. 142. The subchannels 1 and 2 can be reproduced bytwo 8PS-PSKs having the same phase. Returning to FIG. 25(b), as theC-CDM based on the angular shift in the polar coordinate system can beapplied to the PSK as shown in FIG. 141, this will be adopted to thefirst generation satellite broadcast service. However, if adopted to thesecond generation satellite broadcasting based on the APSK, this polarcoordinate system C-CDM is inferior in that signal points in the samegroup cannot be uniformly spaced as shown in FIG. 142. Accordingly,utilization efficiency of electrical power is worsened. On the otherhand, the rectangular coordinate system C-CDM has good compatibility tothe PSK.

The system shown in FIG. 25(b) is compatible with both the rectangularand polar coordinate systems. As the signal points are disposed on theangular lines of the 16 PSK, they can be demodulated by the 16 PSK.Furthermore, as the signal points are divided into groups, the QPSKreceiver can be used for demodulation. Still further, as the signalpoints are also allocated to suit the rectangular coordinate system, thedemodulation will be performed by the 16-SRQAM. Consequently, thecompatibility between the rectangular coordinate system C-CDM and thepolar coordinate system C-CDM can be assured in any of the QPSK, 16PSK,and 16-SRQAM.

The demodulation controller 231 has a memory 231a for storing thereindifferent threshold values (i.e., the shift factors, the number ofsignal points, the synchronization rules, etc.) which correspond todifferent TV broadcast channels. When receiving one of the channelsagain, the values corresponding to the receiving channel will be readout of the memory to thereby stabilize the reception quickly.

If the demodulation data is lost, the demodulation of the second datastream will hardly be executed. This will be explained referring to aflowchart shown in FIG. 24.

Even if the demodulation data is not available, demodulation of the 4PSK at Step 313 and of the first data stream at Step 301 can beimplemented. At Step 302, the demodulation data retrieved by the firstdata stream reproducing unit 232 is transferred to the demodulationcontroller 231. If m is 4 or 2 at Step 303, the demodulation controller231 triggers demodulation of 4 PSK or 2 PSK at Step 313. If not, theprocedure moves to Step 310. At Step 305, two threshold values TH₈ andTH₁₆ are calculated. The threshold value TH₁₆ for AM demodulation is fedat Step 306 from the demodulation controller 231 to both the first 136and the second discrimination/demodulation circuit 137. Hence,demodulation of the modified 16 QAM signal and reproduction of thesecond data stream can be carried out at Steps 307 and 315 respectively.At Step 308, the error rate is examined and if high, the procedurereturns to Step 313 for repeating the 4 PSK demodulation.

As shown in FIG. 22 and the signal points 85, 83, are aligned on a lineat an angle of cos(ωt+nπ/2) while 84 and 86 are off the line. Hence, thefeedback of a second data stream transmitting carrier wave data from thesecond data stream reproducing unit 233 to a carrier reproducing circuit131 is carried out so that no carrier needs to be extracted at thetiming of the signal points 84 and 86.

The transmitter 1 is arranged to transmit carrier timing signals atintervals of a given time with the first data stream for the purpose ofcompensation for no demodulation of the second data stream. The carriertiming signal enables one to identify the signal points 83 and 85 of thefirst data stream regardless of demodulation of the second data stream.Hence, the reproduction of carrier wave can be triggered by thetransmitting of carrier data to the carrier reproducing circuit 131.

A determination then made at Step 304 of the flowchart of FIG. 24 as towhether or not m is 16 upon receipt of such a modified 64 QAM signal asshown in FIG. 23. At Step 310, a determination is also made as towhether or not m is more than 64. If it is determined at Step 311 thatthe received signal has no equal distance signal point constellation,the procedure goes to Step 312. The signal point distance TH₆₄ of themodified 64 QAM signal is calculated from:TH₆₄=(A₁+A₂/2)/(A₁+A₂)

This calculation is equivalent to that of TH₁₆ but its resultantdistance between signal points is smaller.

If the signal point distance in the first sub segment 181 is A₃, thedistance between the first 181 and the second sub segment 182 isexpressed by (A₂−2A₃). Then, the average distance is (A₂−2A₃)/(A₁+A₂)which is designated as d₆₄. when d₆₄ is smaller than T₂ which representsthe signal point discrimination capability of the second receiver 33,any two signal points in the segment will hardly be distinguished fromeach other. This judgment is executed at Step 313. If d₆₄ is out of apermissive range, the procedure moves back to Step 313 for 4 PSK modedemodulation. If d₆₄ is within the range, the procedure advances to Step305 for allowing the demodulation of 16 QAM at Step 307. If it isdetermined at Step 308 that the error rate is too high, the proceduregoes back to Step 313 for 4 PSK mode demodulation.

When the transmitter 1 supplied a modified 8 QAM signal such as shown inFIG. 25(a) in which all the signal points are at angles of cos(2πf+π/4), the carrier waves of the signal are lengthened to the samephase and will thus be reproduced with much ease. At the time, two-bitdata of the first data stream are demodulated by the 4-PSK receiverwhile one-bit data of the second data stream is demodulated by thesecond receiver 33 and the total of three-bit data can be reproduced.

The third receiver 43 will be described in more detail. FIG. 26 shows ablock diagram of the third receiver 43 similar to that of the secondreceiver 33 in FIG. 21. The difference is that a third data streamreproducing unit 234 is added and also, the discrimination/demodulationcircuit has a capability of identifying eight-bit data. The antenna 42of the third receiver 43 has a radius r₃ greater than r₂ thus allowingsmaller distance state signals, e.g. 32- or 64-state QAM signals, to bedemodulated. For demodulation of the 64 QAM signal, the firstdiscrimination/reproduction circuit 136 has to identify 8 digital levelsof the detected signal in which seven different threshold levels areinvolved. As one of the threshold values is zero, three are contained inthe first quadrant.

FIG. 27 shows a space diagram of the signal in which the first quadrantcontains three different threshold values.

As shown in FIG. 27, when the three normalized threshold values are TH1₆₄, TH2 ₆₄, and TH3 ₆₄ they are expressed by:TH1 ₆₄=(A₁+A₃/2)/(A₁+A₂) TH2 ₆₄=(A₁+A₂/2)/(A₁+A₂) andTH3 ₆₄=(A₁+A₂−A₃/2)/(A₁+A₂)

Through AM demodulation of a phase detected signal using the threethreshold values, the third data stream can be reproduced like the firstand second data stream explained with FIG. 21. The third data streamcontains e.g. four signal points 201, 202, 203, and 204 at the first subsegment 181 shown in FIG. 23 which represent 4 values of two-bitpattern. Hence, six digits or modified 64 QAM signals can bedemodulated.

The demodulation controller 231 detects the value m, A₁, A₂, and A₃ fromthe demodulation data contained in the first data stream demodulated bythe first data stream reproducing unit 232 and calculates the threethreshold values TH1 ₆₄, TH2 ₆₄, and TH3 ₆₄ which are then fed to thefirst 136 and the second discrimination/demodulation circuit 137 so thatthe modified 64 QAM signal is demodulated with certainty. Also, if thedemodulation data have been scrambled, the modified 64 QAM signal can bedemodulated only with a specific or subscriber receiver. FIG. 28 is aflowchart showing the action of the demodulation controller 231 formodified 64 QAM signals. The difference from the flowchart fordemodulation of 16 QAM shown in FIG. 24 will be explained. The proceduremoves from Step 304 to Step 320 where it is determined whether or notm=32 or not. If m=32, demodulation of 32 QAM signals is executed at Step322. If not, the procedure moves to Step 321 where it is determinedwhether or not m=64. If yes, A₃ is examined at Step 323. If A₃ issmaller than a predetermined value, the procedure moves to Step 305 andthe same sequence as of FIG. 24 is implemented. If it is judged at Step323 that A₃ is not smaller than the predetermined value, the proceduregoes to Step 324 where the threshold values are calculated. At Step 325,the calculated threshold values are fed to the first and seconddiscrimination/demodulation circuits and at Step 326, the demodulationof the modified 64 QAM signal is carried out. Then, the first, second,and third data streams are reproduced at Step 327. At Step 328, theerror rate is examined. If the error rate is high, the procedure movesto Step 305 where the 16 QAM demodulation is repeated and if low, thedemodulation of the 64 QAM is continued.

The action of carrier wave reproduction needed for execution of asatisfactory demodulating procedure will now be described. The scope ofthe present invention includes reproduction of the first data stream ofa modified 16 or 64 QAM signal using a 4 PSK receiver. However, a common4 PSK receiver rarely reconstructs carrier waves, thus failing toperform a correct demodulation. For compensation, some arrangements arenecessary at both the transmitter and receiver sides.

Two techniques for compensation are provided according to the presentinvention. A first technique relates to transmission of signal pointsaligned at angles of (2n−1)π/4 at intervals of a given time. A secondtechnique offers transmission of signal points arranged at intervals ofan angle of nπ/8.

According to the first technique, the eight signal points including 83and 85 are aligned at angles of π/4, 3π/4, 5π/4, and 7π/4, as shown inFIG. 38. In action, at least one of the eight signal points istransmitted during sync time slot periods 452, 453, 454, and 455arranged at equal intervals of time in a time slot gap 451 shown in thetime chart of FIG. 38. Any desired signal points are transmitted duringthe other time slots. The transmitter 1 is also arranged to assign adata for the time slot interval to the sync timing data region 499 of async data block, as shown in FIG. 41.

The content of a transmitting signal will be explained in more detailreferring to FIG. 41. The time slot group 451 containing the sync timeslots 452, 453, 454, and 455 represents a unit data stream or block 491carrying a data of Dn.

The sync time slots in the signal are arranged at equal intervals of agiven time determined by the time slot interval or sync timing data.Hence, when the arrangement of the sync time slots is detected,reproduction of carrier waves will be executed slot by slot throughextracting the sync timing data from their respective time slots. Such async timing data S is contained in a sync block 493 at the front end ofa data frame 492, which consists of a number of sync time slots denotedby the hatching in FIG. 41. Accordingly, the data to be extracted forcarrier wave reproduction are increased, thus allowing the 4 PSKreceiver to reproduce desired carrier waves at higher accuracy andefficiency.

The sync block 493 comprises sync data regions 496, 497, and 498,—containing sync data S1, S2, and S3, —respectively which include uniquewords and demodulation data. The phase sync signal assignment region 499is at the end of the sync block 493, which holds a data of I_(T)including information about interval arrangement and assignment of thesync time slots.

The signal point data in the phase sync time slot has a particular phaseand can thus be reproduced by the 4 PSK receiver. Accordingly, I_(T) inthe phase sync signal assignment region 499 can be retrieved withouterror thus ensuring the reproduction of carrier waves at with accuracy.

As shown in FIG. 41, the sync block 493 is followed by a demodulationdata block 501 which contains demodulation data about threshold voltagesneeded for demodulation of the modified multiple-bit QAM signal. Thisdata is essential for demodulation of the multiple-bit QAM signal andmay preferably be contained in a region 502 which is a part of the syncblock 493 for ease of retrieval.

FIG. 42 shows the assignment of signal data for transmission of burstform signals through a TDMA method.

The assignment is distinguished from that of FIG. 41 by the fact that aguard period 521 is inserted between any two adjacent Dn data blocks 491and 491 for interruption of the signal transmission. Also, each datablock 491 is at the front end of a sync region 522, the signal points ata phase of (2n−1)π/4 are only transmitted. Accordingly, the carrier wavereproduction will be feasible with the 4 PSK receiver. Morespecifically, the sync signal and carrier waves can be reproducedthrough the TDMA method.

The carrier wave reproduction of the first receiver 23 shown in FIG. 19will be explained in more detail referring to FIGS. 43 and 44. As shownin FIG. 43, an input signal is fed through the input unit 24 to a syncdetector circuit 541 where it is sync detected. A demodulated signalfrom the sync detector 541 is transferred to an output circuit 542 forreproduction of the first data stream. A data of the phase sync signalassignment data region 499 (shown in FIG. 41) is retrieved by anextracting timing controller circuit 543 so that the timing of syncsignals of (2n−1)π/4 data can be acknowledged and transferred as a phasesync control pulse 561 shown in FIG. 44 to a carrier reproductioncontrolling circuit 544. Also, the demodulated signal of the syncdetector circuit 541 is fed to a frequency multiplier circuit 545 whereit is 4× multiplied prior to being transmitted to the carrierreproduction controlling circuit 544. The resultant signal denoted by562 in FIG. 44 contains true phase data 563 and other data. Asillustrated by 564 in the time chart 564 of FIG. 44, the phase sync timeslots 452 carrying the (2n−1)π/4 data are also contained at equalintervals. In the carrier reproducing controlling circuit 544, thesignal 562 is sampled by the phase sync control pulse 561 to produce aphase sample signal 565 which is then converted through a sample andhold operation into a phase signal 566. The phase signal 566 of thecarrier reproduction controlling circuit 544 is fed through a loopfilter 546 to a VCO 547 where its relevant carrier wave is reproduced.The reproduced carrier is then sent to the sync detector circuit 541.

In this manner, the signal point data of the (2n−1)π/4 phase denoted bythe shaded areas in FIG. 39 is recovered and utilized so that a correctcarrier wave can be reproduced by 4× or 16× frequency multiplication.Although a plurality of phases are reproduced at the time, the absolutephases of the carrier can be successfully be identified using a uniqueword assigned to the sync region 496 shown in FIG. 41.

For transmission of a modified 64 QAM signal such as shown in FIG. 40,signal points in the phase sync areas 471 at the (2n−1)π/4 phase denotedby the hatching are assigned to the sync time slots 452, 452b, etc. Itscarrier can hardly be reproduced with a common 4 PSK receiver but can besuccessfully reproduced with the first receiver 23 of 4 PSK modeprovided with the carrier reproducing circuit of the embodiment.

The foregoing carrier reproducing is of COSTAS type. A carrierreproducing circuit of the reverse modulation type will now be explainedaccording to the embodiment.

FIG. 45 shows a reverse modulation type carrier reproducing circuitaccording to the present invention, in which a received signal is fedfrom the input unit 24 to a sync detector circuit 541 for producing ademodulated signal. Also, the input signal is delayed by a first delaycircuit 591 to a delay signal. The delay signal is then transferred to aquadrature phase modulator circuit 592 where it is reverse demodulatedby the demodulated signal from the sync detector circuit 541 to acarrier signal. The carrier signal is fed through a carrier reproductioncontroller circuit 544 to a phase comparator 593. A carrier waveproduced by a VCO 547 is delayed by a second delay circuit 594 into adelay signal which is also fed to the phase comparator 593. At the phasecomparator 593, the reverse demodulated carrier signal is compared inphase with the delay signal thus producing a phase difference signal.The phase difference signal is fed through a loop filter 546 to the VCO547 which in turn produces a carrier wave arranged in phase with thereceived carrier wave. In the same manner as of the COSTAS carrierreproducing circuit shown in FIG. 43, an extracting timing controllercircuit 543 performs sampling of signal points contained in the hatchingareas of FIG. 39. Accordingly, the carrier wave of a 16 or 64 QAM signalcan be reproduced with the 4 PSK demodulator of the first receiver 23.

The reproduction of a carrier wave by 16× frequency multiplication willbe explained. The transmitter 1 shown in FIG. 1 is arranged to modulateand transmit a modified 16 QAM signal with assignment of its signalpoints at nπ/8 phase as shown in FIG. 46. At the first receiver 23 shownin FIG. 19, the carrier wave can be reproduced with its COSTAS carrierreproduction controller circuit containing a 16× multiplier circuit 661shown in FIG. 48. The signal points at each nπ/8 phase shown in FIG. 46are processed at the first quadrant b the action of the 16× multipliercircuit 661, whereby the carrier will be reproduced by the combinationof a loop filter 546 and a VCO 541 547. Also, the absolute phase may bedetermined from 16 different phases by assigning a unique word to thesync region.

The arrangement of the 16× multiplier circuit will be explainedreferring to FIG. 48. A sum signal and a difference signal are producedfrom the demodulated signal by an adder circuit 662 and a subtractorcircuit 663 respectively and then, multiplied together by a multiplier664 into a cos 2θ signal. Also, a multiplier 665 produces a sin 2θsignal. The two signals are then multiplied by a multiplier 666 into asin 4θ signal.

Similarly, a sin 8θ signal is produced from the two, sin 2θ and cos 2θ,signals by the combination of an adder circuit 667, a subtracter circuit668, and a multiplier 670. Furthermore, a sin 16θ signal is produced bythe combination of an adder circuit 671, a subtractor circuit 672, and amultiplier 673. Then, the 16× multiplication is completed.

Through the foregoing 16× multiplication, the carrier wave of all thesignal points of the modified 16 QAM signal shown in FIG. 46 willsuccessfully be reproduced without extracting particular signal points.

However, reproduction of the carrier wave of the modified 64 QAM signalshown in FIG. 47 can involve an increase in the error rate due todislocation of some signal points from the sync areas 471.

Two techniques are known for compensation for the consequences. One isinhibiting transmission of the signal points dislocated from the syncarea. This causes the total amount of transmitted data to be reduced butallows the arrangement to be facilitated. The other is providing thesync time slots as described in FIG. 38. In more particular, the signalpoints in the nπ/8 sync phase areas, e.g. 471 and 471a, are transmittedduring the period of the corresponding sync time slots in the time slotgroup 451. This triggers an accurate synchronizing action during theperiod thus minimizing phase error.

As now understood, the 16× multiplication allows the simple 4 PSKreceiver to reproduce the carrier wave of a modified 16 or 64 QAMsignal. Also, the insertion of the sync time slots causes the phasicaccuracy to be increased during the reproduction of carrier waves from amodified 64 QAM signal.

As set forth above, the signal transmission system of the presentinvention is capable of transmitting a plurality of data on a singlecarrier wave simultaneously in the multiple signal level arrangement.

More specifically, three different level receivers which have discretecharacteristics of signal intercepting sensitivity and demodulatingcapability are provided in relation to one single transmitter so thatany one of them can be selected depending on a wanted data size to bedemodulated which is proportional to the price. When the first receiverof low resolution quality and low price is acquired together with asmall antenna, its owner can intercept and reproduce the first datastream of a transmission signal. When the second receiver of mediumresolution quality and medium price is acquired together with a mediumantenna, its owner can intercept and reproduce both the first and seconddata streams of the signal. When the third receiver of high resolutionquality and high price is acquired with a large antenna, its owner canintercept and reproduce all the first, second, and third data streams ofthe signal.

If the first receiver is a home-use digital satellite broadcast receiverof low price, it will overwhelmingly be welcome by a majority ofviewers. The second receiver accompanied with the medium antenna costsmore and will be accepted by not common viewers but particular peoplewho want to enjoy HDTV services. The third receiver accompanied with thelarge antenna at least before the satellite output is increased, is notappropriate for home use and will possibly be used in relevantindustries. For example, the third data stream carrying super HDTVsignals is transmitted via a satellite to subscriber cinemas which canthus play video tapes rather than traditional movie films and run moviesat low cost.

When the present invention is applied to a TV signal transmissionservice, three different quality pictures are carried on one signalchannel wave and will offer compatibility with each other. Although thefirst embodiment refers to a 4 PSK, a modified 8 QAM, a modified 16 QAM,and a modified 64 QAM signal, other signals will also be employed withequal success including a 32 QAM, a 256 QAM, an 8 PSK, and a 16 PSK, anda 32 PSK signal. It would be understood that the present invention isnot limited to a satellite transmission system and can be applied to aterrestrial communications system or a cable transmission system.

The transmission method of the invention can also be applied to a4-level or 8-level ASK signal as shown in FIG. 58 and FIGS. 68(a) and(b), respectively.

EMBODIMENT 2

A second embodiment of the present invention is featured in which thephysical multi-level arrangement of the first embodiment is divided intosmall levels through e.g. discrimination in error correction capability,thus forming a logic multi-level construction. In the first embodiment,each multi-level channel has different levels in the electrical electricsignal amplitude or physical demodulating capability. The secondembodiment offers different levels in the logic reproduction capabilitysuch as error correction. For example, the data D₁ in a multi-levelchannel is divided into two, D₁₋₁ and D₁₋₂, components and D₁₋₁ is moreincreased in the error correction capability than D₁₋₂ fordiscrimination. Accordingly, as the error detection and correctioncapability is different between D₁₋₁ and D₁₋₂ at demodulation, D₁₋₁ cansuccessfully be reproduced within a given error rate when the C/N levelof an original transmitting signal is as low as disenabling thereproduction of D₁₋₂. This will be implemented using the logicmulti-level arrangement.

More specifically, the logic multi-level arrangement consists ofdividing data of a modulated multi-level channel and discriminatingdistances between error correction codes by mixing error correctioncodes with product codes for varying error correction capability. Hence,a more multi-level signal can be transmitted.

In fact, a D₁₋₁ channel is divided into two sub channels D₁ and D₁₋₂ anda D₂ channel is divided into two sub channels D₂₋₁ and D₂₋₂.

This will be explained in more detail referring to FIG. 87 85 in whichD₁₋₁ is reproduced from a lowest C/N signal. If the C/N rate is d atminimum, three components D₁₋₂, D₂₋₁ and D₂₋₂ cannot be reproduced whileD₁₋₁ is reproduced. If C/N is not less than c, D₁₋₂ can also bereproduced. Equally, when C/N is b, D₂₋₁ is reproduced and when C/N isa, D₂₋₂ is reproduced. As the C/N rate increases, the reproduciblesignal levels are increased in number. The lower the C/N, the fewer thereproducible signal levels. This will be explained in the form ofrelationship between transmitting distance and reproducible C/N valuereferring to FIG. 86. In common, the C/N value of a received signal isdecreased in proportion to the distance of transmission as expressed bythe real line 861 in FIG. 86. It is now assumed that the distance from atransmitter antenna to a receiver antenna is La when C/N=a, Lb whenC/N=b, Lc when C/N=c, Ld when C/N=d, and Le when C/N=e. If the distancefrom the transmitter antenna is greater than Ld, D₁₋₁ can be reproducedas shown in FIG. 85 where the receivable area 862 is denoted by thehatching. In other words, D₁₋₁ can be reproduced within a most extendedarea. Similarly, D₁₋₂ can be reproduced in an area 863 when the distanceis not more than Lc. In this area 863 containing the area 862, D₁₋₁ canwith no doubt be reproduced. In a small area 854 864, D₂₋₁ can bereproduced and in a smallest area 865, D₂₋₂ can be reproduced. Asunderstood, the different data levels of a channel can be reproducedcorresponding to degrees of declination in the C/N rate. The logicmulti-level arrangement of the signal transmission system of the presentinvention can provide the same effect as of a traditional analoguetransmission system in which the amount of receivable data is graduallylowered as the C/N rate decreases.

The construction of the logic multi-level arrangement will be describedin which there are provided two physical levels and two logic levels.FIG. 87 is a block diagram of a transmitter 1 which is substantiallyidentical in construction to that shown in FIG. 2 and describedpreviously in the first embodiment and will not be further explained indetail. The only difference is that error correction code encoders areadded as abbreviated to ECC encoders. The divider circuit 3 has fouroutputs 1-1, 1-2, 2-1, and 2-2 through which four signals D₁₋₁, D₁₋₂,D₂₋₁, and D₂₋₂ divided from an input signal are delivered. The twosignals D₁₋₁ and D₁₋₂ are fed to two, main and sub, ECC encoders 872aand 873a of a first ECC encoder 871a respectively for converting toerror correction code forms.

The main ECC encoder 872a has a higher error correction capability thanthat of the sub ECC encoder 873a. Hence, D₁₋₁ can be reproduced at alower rate of C/N than D₁₋₂ as apparent from the CN-level diagram ofFIG. 85. More particularly, the logic level of D₁₋₁ is less affected bydeclination of the C/N than that of D₁₋₂. After error correction codeencoding, D₁₋₁ and D₂₋₂ D₁₋₂ are summed by a summer 874a to a D₁ signalwhich is then transferred to the modulator 4. The other two signals D₂₋₁and D₂₋₂ of the divider circuit 3 are error correction encoded by two,main and sub, ECC encoders 872 b 872b and 873b of a second ECC encoder871b respectively and then, summed by a summer 874b to a D₂ signal whichis transmitted to the modulator 4. The main ECC encoder 872b is higherin the error correction capability than the sub ECC encoder 873b. Themodulator 4 in turn produces from the two, D₁ and D₂, input signals amulti-level modulated signal which is further transmitted from thetransmitter unit 5. As understood, the output signal from thetransmitter 1 has two physical levels D₁ and D₂ and also, four logiclevels D₁₋₁, D₁₋₂, D₂₋₁, and D₂₋₂ based on the two physical levels forproviding different error correction capabilities.

The reception of such a multi-level signal will be expelained explained.FIG. 88 is a block diagram of a second receiver 33 which is almostidentical in construction to that shown in FIG. 21 and described in thefirst embodiment. The second receiver 33 arranged for interceptingmulti-level signals from the transmitter 1 shown in FIG. 87 furthercomprises first and second ECC decoder 876a 876b, in which thedemodulation of QAM, or any of ASK, PSK, and FSK if desired, isexecuted.

As shown in FIG. 88, a receiver signal is demodulated by the demodulator35 to the two, D₁ and D₂, signals which are then fed to two dividers 3aand 3b respectively where they are divided into four logic levels D₁₋₁,D₁₋₂, D₂₋₁, and D₂₋₂. The four signals are transferred to the first andsecond ECC decoders 876a and 876b in which D₁₋₁ is error corrected by amain ECC decoder 877a, D₁₋₂ by a sub ECC decoder 878a, D₂₋₁ by a mainECC decoder 877b, D₂₋₂ by a sub ECC decoder 878b before all being sentto the summer 37. In the mixer 37, the four, D₁₋₁, D₁₋₂, D₂₋₁, and D₂₋₂,error corrected signals are combined into a signal which is thendelivered from the output unit 36.

Since D₁₋₁ and D₂₋₂ are higher in the error correction capability thanD₁₋₂ and D₂₋₂ respectively, the error rate remains less than a givenvalue although C/N is fairly low as shown in FIG. 85 and thus, anoriginal signal will be reproduced successfully.

The action of discriminating the error correction capability between themain ECC decoders 877a and 877b of high code gain and the sub ECCdecoders 878a and 878b of low code gain will now be described in moredetail. It is a good idea for having a difference in the errorcorrection capability, i.e., in the code gain, to use in the sub ECCdecoder a common coding technique, e.g. Reed-Solomon or BCH method, asshown in FIG. 165(b) for the ECC decoder, having a standard codedistance and in the the main ECC decoder, another encoding technique inwhich distance between correction codes is increased using Reed-Solomoncodes, their product codes, or other long-length codes or a trellisdecoder 744p, 744q, and 744r shown in FIGS. 128(d), 128(e), 128(f). Avariety of known techniques for increasing the error correction codedistance have been introduced and will not be explained in detail. Thepresent invention can be associated with any known technique for havingthe logic multi-level arrangement.

Also, as shown in the block diagram of FIGS. 160 and 167, thetransmitter further has an interleaver 744k and the receiver further hasde-interleavers 759k and 936b. The interleave process is carried out bythe use of the Interleave Table 954 shown in FIG. 168(a). De-interleaveRAM 936x in the de-interleaver 936b is used for decoding the data. Bythis arrangement, the data transmission system having high reliabilitywith respect to the burst error can be realized, resulting in stabletransmitted images.

The logic multi-level arrangement will be explained in conjuctionconjunction with a diagram of FIG. 89 showing the relationship betweenC/N and error race rate after error correction. As shown, the straightline 881 represents D₁₋₁ at the C/N and error rate relation and the line882 represents D₁₋₂ at same.

As the C/N rate of an input signal decreases, the error rate increasesafter error correction. If C/N is lower than a given value, the errorrate exceeds a reference value Eth determined by the system designstandards and no original data will normally be reconstructed. When C/Nis lowered to less than e, the D₁ signal fails to be reproduced asexpressed by the line 881 of D₁₋₁ in FIG. 89. When e≦C/N<d, D₁₋₁ of theD₁ signal exhibits a higher error rate than Eth and will not bereproduced.

When C/N is d at the point 885d, D₁₋₁ having a higher error correctioncapability than D₁₋₂ becomes not higher in the error rate than Eth andcan be reproduced. At the time, the error rate of D₁₋₂ remains higherthan Eth after error correction and will no longer be reproduced.

When C/N is increased up to c at the point 885c, D₁₋₂ becomes not higherin the error rate than Eth and can be reproduced. At the time, D₂₋₁ andD₂₋₂ remain in no demodulation state. After the C/N rate is increasedfurther to b′, the D₂ signal becomes ready to be demodulated.

When C/N is increased to b at the point 885b, D₂₋₁ of the D₂ signalbecomes not higher in the error rate than Eth and can be reproduced. Atthe time, the error rate of D₂₋₂ remains higher than Eth and will not bereproduced. When C/N is increased up to a at the point 885a, D₂₋₂becomes not higher than Eth and can be reproduced.

As described above, the four different signal logic levels divided fromtwo, D₁ and D₂, physical levels through discrimination of the errorcorrection capability between the levels, can be transmittedsimultaneously.

Using the logic multi-level arrangement of the present invention with amulti-level construction in which at least a part of the original signalis reproduced even if data in a higher level is lost, digital signaltransmission will successfully be executed without losing theadvantageous effect of an analogue signal transmission in whichtransmitting data is gradually decreased as the C/N rate becomes low.

ThankingThanks to up-to-date compression techniques, compressed imagedata can be transmitted in the logic multi-level arrangement forenabling a receiver station to reproduce a higher quality image thanthat of an analogue system and also, with not sharply but at stepsdeclining the signal level for ensuring signal interception in a widerarea. The present invention can provide an extra effect of themulti-layer arrangement which is hardly implemented by a known digitalsignal transmission system without deteriorating high quality imagedata.

In addition, the address data of the image segment data, the base imagedata for image compression, the scramble cancellation data shown in thedescrambler (FIG. 66), and high priority (HP) data, i.e., the data(e.g., the frame synchronization signal and header) that is mostessential to image expansion of the HDTV signal, is transmitted as D₁₋₁by the high code gain ECC encoder 743a (FIGS. 88, 133, 170, and 172),and is received by the high code gain ECC decoder 758 of the receiver43.

This high priority data is protected because the error rate of prioritydata D₁₋₁ does not increase noticeably. Fatal deterioration of thecharacteristic image quality of digital video transmissions is thusavoided, and a “graceful degradation” effect whereby image qualitygradually deteriorates is obtained. The modulator 749 and demodulator760 of FIGS. 133 and 170, respectively, can achieve this gracefuldegradation effect with 16-level QAM and 32-level QAM described above,4-level VSB (FIG. 57) and 8-level VSB (FIG. 68) described below in thedescription of the fourth embodiment, and 8-level PSK.

Furthermore, as shown in the block diagrams of FIGS. 133 and 156, a bigdifference in the error rate of high priority data and low priority datacan be created during signal reception by applying high code gain errorcorrection coding of the high priority data by means of the ECC encoder744a and trellis encoder 744b in the 2nd data stream input 744, whileerror correction encoding the low priority data with low code gain bythe ECC encoder 743a only.

As a result, even if the C/N ratio of the transmission systemdeteriorates significantly, the high priority data can be received.Therefore, while the image quality deteriorates with the deteriorationof the low priority data, the high priority data can also be reproducedin applications subject to severe C/N ratio deterioration, as found inthe reception conditions encountered with mobile television receivers,and the pixel block positioning information is also reproduced. Becauseimage block destruction is thus prevented, viewers are still able toreceive and view broadcast programming under extremely poor receptionconditions.

EMBODIMENT 3

A third embodiment of the present invention will be described referringto the relevant drawings.

FIG. 29 is a schematic total view illustrating the third embodiment inthe form of a digital TV broadcasting system. An input video signal 402of super high resolution TV image is fed to an input unit 403 of a firstvideo encoder 401. Then, the signal is divided by a divider circuit 404into three, first, second, and third, data streams which are transmittedto a compressing circuit 405 for data compression before being furtherdelivered.

Equally, other three input video signals 406, 407, and 408 are fed to asecond 409, a third 410, and a fourth video encoder 411 respectivelywhich all are arranged identical in construction to the first videoencoder 401 for data compression.

The four first data streams from their respective encoders 401, 409,410, and 411 are transferred to a first multiplexer 413 of a multiplexer412 where they are time multiplexed by a TDM process into a first datastream multiplex signal which is fed to a transmitter 1.

A part or all of the four second data streams from their respectiveencoders 401, 409, 410, and 411 are transferred to a second multiplexer414 of the multiplexer 412 where they are time multiplexed to a seconddata stream multiplex signal which is then fed to transmitter 1. Also, apart or all of the four third data streams are transferred to a thirdmultiplexer 415 where they are time multiplexed to a data streammultiplex signal which is then fed to the transmitter 1.

The transmitter 1 performs modulation of the three data stream signalswith its modulator 4 by the same manner as described in the firstembodiment. The modulated signals are sent from a transmitter unit 5through an antenna 6 and an uplink 7 to a transponder 12 of a satellite10 which in turn transmits it to three different receivers including afirst receiver 23.

The modulated signal transmitted through a downlink 21 is intercepted bya small antenna 22 having a radius r₁ and fed to a first data streamreproducing unit 232 of the first receiver 23 where its first datastream only is demodulated. The demodulated first data stream is thenconverted by a first video decoder 421 to a traditional 425 orwide-picture NTSC or video output signal 426 of low image resolution.

Also, the modulated signal transmitted through a downlink 31 isintercepted by a medium antenna 32 having a radius r₂ and fed to a first232 and a second data stream reproducing unit 233 of a second receiver33 where its first and second data streams are demodulated respectively.The demodulated first and second data streams are then summed andconverted by a second video decoder 422 to an HDTV or video outputsignal 427 of high image resolution and/or to the video output signals425 and 426.

Also, the modulated signal transmitted through a downlink 41 isintercepted by a large antenna 42 having a radius r₃ and fed to a first232, a second 233, and a third data steam reproducing unit 234 of athird receiver 43 where its first, second, and third data streams aredemodulated respectively. The demodulated first, second, and third datastreams are then summed and converted by a third video decoder 423 to asuper HDTV or video output signal 428 of super high image resolution foruse in a video theater or cinema. The video output signals 425, 426, and427 can also be reproduced if desired. A common digital TV signal istransmitted from a conventional digital transmitter 51 and whenintercepted by the first-receiver 23, will be converted to the videooutput signal 426 such as a low resolution NTSC TV signal.

The first video encoder 401 will now be explained in more detailreferring to the block diagram of FIG. 30. An input video signal ofsuper high resolution is fed through the input unit 403 to the dividercircuit 404 where it is divided into four components by sub-band codingprocess. In particular, the input video signal is separated by passingthrough a horizontal lowpass filter 451 and a horizontal highpass filter452 of e.g. QAM mode to two, low and high, horizontal frequencycomponents which are then subsampled into half of their quantities bytwo subsamplers 453 and 454 respectively. The low horizontal componentis filtered by a vertical lowpass filter 455 and a vertical highpassfilter 456 into a low horizontal low vertical component or H_(L)V_(L)signal and a low horizontal high vertical component or H_(L)V_(H) signalrespectively. The two, H_(L)V_(L) and H_(L)V_(H), signals are thensubsampled into one half by two subsamblers 457 and 458 respectively andtransferred to the compressing circuit 405.

The high horizontal component is filtered by a vertical lowpass filter459 and a vertical highpass filter 460 into a high horizontal lowvertical highpass component or H_(H)V_(L) signal and a high horizontalhigh vertical component or H_(H)V_(H) signal respectively. The two,H_(H)V_(L) and H_(H)V_(H), signals are then subsampled into one half bytwo subsamplers 461 and 462 respectively and transferred to thecompressing circuit 405.

The H_(L)V_(L) signal is preferably DCT compressed by a first compressor471 of the compressing circuit 405 and fed to a first output circuit 472as the first data stream.

Also, the H_(L)V_(H) signal is compressed by a second compressor 473 andfed to a second output circuit 464. The H_(H)V_(L) signal is compressedby a third compressor 463 and fed to the second output circuit 464.

The H_(H)V_(H) signal is divided by a divider 465 into two highresolution (H_(H)V_(H)1) and super high resolution (H_(H)V_(H)2) videosignals which are then transferred to the second output circuit 464 anda third output circuit 468 respectively.

The first video decoder 421 will now be explained in more detailreferring to FIG. 31. The first data stream or D₁ signal of the firstreceiver 23 is fed through an input unit 501 to a descrambler 502 of thefirst video decoder 421 where it is descrambled. The descrambled D₁signal is expanded by an expander 503 to H_(L)V_(L) which is then fed toan aspect ratio changing circuit 504. Thus, the H_(L)V_(L) signal can bedelivered through an output unit 505 as a standard 500, letterbox format507, wide-screen 508, or sidepanel format NTSC signal 509. The scanningformat may be of non-interlace or interlace type and its NTSC mode linesmay be 525 or doubled to 1050 by double tracing. When the receivedsignal from the digital transmitter 51 is a digital TV signal of 4 PSKmode, it can also be converted by the first receiver 23 and the firstvideo decoder 421 to a TV picture. The second video decoder 422 will beexplained in more detail referring to the block diagram of FIG. 32. TheD₁ signal of the second receiver 33 is fed through a first input 521 toa first expander 522 for data expansion and then, transferred to anoversampler 523 where it is sampled at 2x. The oversampled signal isfiltered by a vertical lowpass filter 524 into H_(L)V_(L). Also, the D₂signal of the second receiver 33 is fed through a second input 530 to adivider 531 where it is divided into three components which are thentransferred to second, third, and fourth expanders 532-534 respectivelyfor data expansion. The three expanded components are sampled at 2x bythree oversamplers 535, 536, and 537 and filtered by a vertical highpass538, a vertical lowpass 539, and a vertical highpass filter 540respectively. Then, H_(L)V_(L) from the vertical lowpass filter 524 andH_(L)V_(H) from the vertical highpass filter 538 are summed by an adder525, sampled by an oversampler 541, and filtered by a horizontal lowpassfilter 542 into a low frequency horizontal video signal. H_(H)V_(L) fromthe vertical lowpass filter 539 and H_(H)V_(H)1 from the verticalhighpass filter 540 are summed by an adder 526, sampled by anoversampler 544, and filtered by a horizontal highpass filter 545 to ahigh frequency horizontal video signal. The two, high and low frequency,horizontal video signals are then summed by an adder 543 into a highresolution video signal HD which is further transmitted through anoutput unit 546 as a video output 547 of e.g. HDTV format. If desired atraditional NTSC video output can be reconstructed with equal success.

FIG. 33 is a block diagram of the third video decoder 423 in which theD₁ and D₂ signals are fed through a first 521 and a second input 530respectively to a high frequency band video decoder circuit 527 wherethey are converted to an HD signal in the same manner as describedabove. The D₃ signal is fed through a third input 551 to a super highfrequency band video decoder circuit 552 where it is expanded,descrambled, and composed into H_(H)V_(H)2 signal. The HD signal of thehigh frequency band video decoder circuit 527 and the H_(H)V_(H)2 signalof the super high frequency band video decoder circuit 552 are summed bya summer 553 to a super high resolution TV or S-HD signal which is thendelivered through an output unit 554 as a super resolution video output555.

The action of multiplexing in the multiplexer 412 shown in FIG. 29 willbe explained in more detail. FIG. 34 illustrates a data assignment inwhich the three, first, second, and third, data streams D₁, D₂, D₃contain in a period of T six NTSC channel data L1, L2, L3, L4, L5, L6,six HDTV channel data M1, M2, M3, M4, M5, M6 and six S-HDTV channel dataH1, H2, H3, H4, H5, H6 respectively. In operation, the NTSC or D₁ signaldata L1 to L6 are time multiplexed by TDM process during the period T.More particularly, H_(L)V_(L) of D₁ is assigned to a domain 601 for thefirst channel. Then, a difference data M1 between HDTV and NTSC or a sumof H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H)1 is assigned to a domain 602for the first channel. Also, a difference data HI H1 between HDTV andsuper HDTV or H_(H)V_(H)2 (See FIG. 30) is assigned to a domain 603 forthe first channel.

The selection of the first channel TV signal will now be described. Whenintercepted by the first receiver 23 with a small antenna coupled to thefirst video decoder 421, the first channel signal is converted to astandard or widescreen NTSC TV signal as shown in FIG. 31. Whenintercepted by the second receiver 33 with a medium antenna coupled tothe second video decoder 422, the signal is converted by summing L1 ofthe first data stream D₁ assigned into the domain 601 and M1 of thesecond data stream D₂ assigned to the domain 602 to an HDTV signal ofthe first channel equivalent in program to the NTSC signal.

When intercepted by the third receiver 43 with a large antenna coupledto the third video decoder 423, the signal is converted by summing L1 ofD₁ assigned to the domain 601, M1 of D₂ assigned to the domain 602, andH₁ of D₃ assigned to the domain 603 into a super HDTV signal of thefirst channel equivalent in program to the NTSC signal. The otherchannel signals can be reproduced in an equal manner.

FIG. 35 shows another data assignment L1 of a first channel NTSC signalis assigned to a fistfirst domain 601. The domain 601 which is allocatedat the front end of the first data stream D₁, also contains at front adata S₁₁ including a descrambling data and the demodulation datadescribed in the first embodiment. A first channel HDTV signal istransmitted as L1 and M1. M1, which is thus a difference data betweenNTSC and HDTV, is assigned to two domains 602 and 611 of D₂. If L1 is acompressed NTSC component of 6 Mbps, M1 is two times higher, that is, 12Mbps. Hence, the total of L1 and M1 can be demodulated at 18 Mbps withthe second receiver 33 and the second video decoder 423. According tocurrent data compression techniques, HDTV compressed signals can bereproduced at about 15 Mbps. This allows the data assignment shown inFIG. 35 to enable simultaneous reproduction of an NTSC and HDTV firstchannel signal. However, this assignment allows no second channel HDTVsignal to be carried. S21 is a descrambling data in the HDTV signal. Afirst channel super HDTV signal component comprises L1, M1, and H1. Thedifference data H1 is assigned to three domains 603, 612, and 613 of D₃.If the NTSC signal is 6 Mbps, the super HDTV is as high as 36 Mbps. Whena compressed rate is increased, super HDTV video data of about 2000scanning line for reproduction of a cinema size picture for commercialuse can be transmitted in an equal manner.

FIG. 36 shows a further data assignment in which H1 of a super HDTVsignal is assigned to six time domains. If a NTSC compressed signal is 6Mbps, this assignment can be nine times higher, that is, 54 Mbps of D₃data. Accordingly, super HDTV data of higher picture quality can betransmitted.

The foregoing data assignment makes the use of one of two, horizontaland vertical, polarization planes of a transmission wave. When both thehorizontal and vertical polarization planes are used, the frequencyutilization will be doubled. This will be explained below.

FIG. 49 shows a data assignment in which D_(V1) and D_(H1) are avertical and a horizontal polarization signal of the first data streamrespectively, D_(V2) and D_(H2) are a vertical and a horizontalpolarization signal of the second data stream respectively, and D_(V3)and D_(H3) are a vertical and a horizontal polarization signal of thethird data stream respectively. The vertical polarization signal D_(V1)of the first data stream carries a low frequency band or NTSC TV dataand the horizontal polarization signal D_(H1) carries a high frequencyband or HDTV data. When the first receiver 23 is equipped with avertical polarization signal D_(H1) carries a high frequency band orHDTV data. When the first receiver 23 is equipped with an antenna forboth horizontally and vertically polarized waves, it can reproduce theHDTV signal through summing L1 and M1. More specifically, the firstreceiver 23 can provide compatibility between NTSC and HDTV with the useof a particular type antenna.

FIG. 50 illustrates a TDMA method in which each data burst 721 isaccompanied at front a sync data 731 and a card data 741. Also, a framesync data 720 is provided at the front of a frame. Like channels areassigned to like time slots. For example, a first time slot 750 carriesNTSC, HDTV, and super HDTV data of the first channel simultaneously. Thesix time slots 750, 750a, 750b, 750c, 750d, 750e, are arrangedindependent from each other. Hence, each station can offer NTSC, HDTV,and/or super HDTV services independently of the other stations throughselecting a particular channel of the time slots. Also, the firstreceiver 23 can reproduce an NTSC signal when equipped with a horizontalpolarization antenna and both NTSC and HDTV signals when equipped with acompatible polarization antenna. In this respect, the second receiver 33can reproduce a super HDTV at lower resolution while the third receiver43 can reproduce a full super HDTV signal. According to the thirdembodiment, a compatible signal transmission system will be constructed.It is understood that the data assignment is not limited to the burstmode TDMA method shown in FIG. 50 and another method such as timedivision multiplexing of continuous signals as shown in FIG. 49 will beemployed with equal success. Also, a data assignment shown in FIG. 51will permit a HDTV signal to be reproduced at high resolution.

As set forth above, the compatible digital TV signal transmission systemof the third embodiment can offer three, super HDTV, HDTV, andconventional NTSC, TV broadcast services simultaneously. In addition, avideo signal intercepted by a commercial station or cinema can beelectronized.

The modified QAM of the embodiments is now termed as SRQAM and its errorrate will be examined.

First, the error rate in 16 SRQAM will be calculated. FIG. 99 shows avector diagram of 16 SRQAM signal points. As apparent from the firstquadrant, the 16 signal points of standard 16 QAM including 83a, 83b 85,84a, 83a 86a are allocated at equal intervals of 2δ.

The signal point 83a is spaced δ from both the I-axis and the Q-axis ofthe coordinate. It is now assumed that n is a shift value of the 16SRQAM. In 16 SRQAM, the signal point 83a of 16 QAM is shifted to asignal point 83 where the distance from each axis is nδ. The shift valuen is thus expressed as:0<n<3.

The other signal points 84a and 86a are also shifted to two points 84and 86 respectively.

If the error rate of the first data stream is Pe₁, it is obtained from:${Pe}_{1 - 16} = {{{\frac{1}{4}{{erfc}\left( \frac{n\delta}{\sqrt{2\sigma}} \right)}} + {\frac{1}{4}{{erfc}\left( \frac{3\delta}{\sqrt{2\sigma}} \right)}}}\quad = {\frac{1}{8}{{erfc}\left( \frac{n\sqrt{p}}{\sqrt{9 + n^{2}}} \right)}}}$

Also, the error rate Pe₂ of the second data stream is obtained from:$\begin{matrix}{{Pe}_{2 - 16} = {\frac{1}{2}{{erfc}\left( \frac{\frac{3 - n}{2}\delta}{\sqrt{2\sigma}} \right)}}} \\{= {\frac{1}{4}{{erfc}\left( {\frac{\frac{3 - n}{2}\delta}{2\sqrt{9 + n^{2}}}\sqrt{p}} \right)}}}\end{matrix}$

The error rate of 36 or 32 SRQAM will be calculated. FIG. 100 is avector diagram of a 36 SRQAM signal in which the distance between anytwo 36 QAM signal points is 2δ.

The signal point 83a of 36 QAM is spaced δ from each axis of thecoordinate. It is now assumed that n is a shift value of the 16 SRQAM.In 36 SRQAM, the signal point 83a is shifted to a signal point 83 wherethe distance from each axis is nδ. Similarly, the nine 36 QAM signalpoints in the first quadrant are shifted to points 83, 84, 85, 86, 97,98, 99, 100, 101 respectively. If a signal point group 90 comprising thenine signal points is regarded as a single signal point, the error ratePe₁ in reproduction of only the first data stream D₁ with a modified 4PSK receiver and the error rate Pe₂ in reproduction of the second datastream D₂ after discriminating the nine signal points of the group 90from each other, are obtained respectively from: $\begin{matrix}{{Pe}_{1 - 32} = {\frac{1}{6}{{erfc}\left( \frac{n\delta}{\sqrt{2\sigma}} \right)}}} \\{= {\frac{1}{6}{{erfc}\left( {\sqrt{\frac{6p}{5}} \times \frac{n}{\sqrt{n^{2} + {2n} + 25}}} \right)}}}\end{matrix}$ $\begin{matrix}{{Pe}_{2 - 32} = {\frac{2}{3}{{erfc}\left( {\frac{5 - n}{4\sqrt{22}}\frac{\delta}{p}} \right)}}} \\{= {\frac{2}{3}{{erfc}\left( {\sqrt{\frac{3p}{40}} \times \frac{5 - n}{\sqrt{n^{2} + {2n} + 25}}} \right)}}}\end{matrix}$

FIG. 101 shows the relationship between error rate Pe and C/N rate intransmission in which the curve 900 represents a conventional or notmodified 32 QAM signal. The straight line 905 represents a signal having10^(−1.5) of the error rate. The curve 901a represents a D₁ level 32SRQAM signal of the present invention at the shift rate n of 1.5. Asshown, the C/N rate of the 32 SRQAM signal is 5 dB lower at the errorrate of 10^(−1.5) than that of the conventional 32 QAM. This means thatthe present invention allows a D₁ signal to be reproduced at a givenerror rate when its C/N rate is relatively low.

The curve 902a represents a D₂ level SRQAM signal at n=1.5 which can bereproduced at the error rate of 10^(−1.5) only when its C/N rate is 2.5dB higher than that of the conventional 32 QAM of the curve 900. Also,the curves 901b and 902b represent D₁ and D₂ SRQAM signals at n=2.0respectively. The curves curve 902c represents a D₂ SRQAM signal atn=2.5. It is apparent that the C/N rate of the SRQAM signal at the errordata of 10^(−1.5) is 5 dB, 8 dB, and 10 dB higher at n=1.5, 2.0, and 2.5respectively in the D₁ level and 2.5 dB lower in the D₂ level than thatof a common 32 QAM signal.

Shown in FIG. 103 is the C/N rate of the first and second data streamsD₁, D₂ of a 32 SRQAM signal which is needed for maintaining a constanterror rate against variation of the shift n. As apparent, when the shiftn is more than 0.8, there is developed a clear difference between twoC/N rates of their respective D₁ and D₂ levels so that the multi-levelsignal, namely first and second data, transmission can be implementedsuccessfully. In brief, n>0.85 is essential for multi-level datatransmission of the 32 SRQAM signal of the present invention.

FIG. 102 shows the relationship between the C/N rate and the error ratefor 16 SRQAM signals. The curve 900 represents a common 16 QAM signal.The curves 901a, 901b, 901c and D₁ level or first data stream 16 SRQAMsignals at n=1.2, 1.5, and 1.8 respectively. The curves 902a, 902b, 902care D₂ level or second data stream 16 SRQAM signals at n=1.2, 1.5, and1.8 respectively.

The C/N rate of the first and second data streams D₁, D₂ of a 16 SRQAMsignal is shown in FIG. 104, which is needed for maintaining a constanterror rate against variation of the shift n. As apparent, when the shiftn is more than 0.9 (n>0.9), the multi-level data transmission of the 16SRQAM signal will be executed.

One example of propagation of SRQAM signals of the present inventionwill now be described for use with a digital TV terrestrial broadcastservice. FIG. 105 shows the relationship between the signal level andthe distance between a transmitter antenna and a receiver antenna in theterrestrial broadcast service. The curve 911 represents a transmittedsignal from the transmitter antenna which is 1250 feet high. It isassumed that the error rate essential for reproduction of an applicabledigital TV signal is 10^(−1.5). The hatching area 912 represents a noiseinterruption. The point 910 represents a signal reception limit of aconventional 32 QAM signal at C/N=15 dB where the distance L is 60 milesand a digital HDTV signal can be intercepted at minimum.

The C/N rate varies 5 dB under a worst case receiving condition such asbad weather. If a change in the relevant condition, e.g. weather,attenuates the C/N rate, the interception of an HDTV signal will hardlybe ensured. Also, geographical conditions largely affect the propagationof signals and a decrease of about 10 dB at least will be unavoidable.Hence, successful signal interception within 60 miles will never beguaranteed and above all, a digital signal will be harder to propagatethan an analogue signal. It would be understood that the service area ofa conventional digital TV broadcast service is less dependable.

In case of the 32 SRQAM signal of the present invention or the 8-VSBshown in FIG. 68, a three-level signal transmission system isconstituted as shown in FIGS. 133 and 137. This permits a low resolutionNTSC signal of MPEG level to be carried on the 1-1 data stream D₁₋₁, amedium resolution TV data of e.g. NTSC system to be carried on the 1-2data stream D₁₋₂, and a high frequency component of HDTV data to becarried on the second data stream D₂. Accordingly, the service area ofthe 1-2 data stream of the SRQAM signal is increased to a 70 mile point910a while that of the second data stream remains within a 55 mile point910b, as shown in FIG. 105. FIG. 106 illustrates a computer simulationresult of the service area of the 32 SRQAM signal of the presentinvention, which is similar to FIG. 53 but explains it in more detail.As shown, the regions 708, 703c, 703a, 703b, and 712 represent aconventional 32 QAM receivable area, a 1-1 data level D₁₋₁ receivablearea, a 1-2 data level D₁₋₂ receivable area, a second data level D₂receivable area, and a service area of a neighbor analogue TV stationrespectively. The conventional 32 QAM signal data used in this drawingis based on a conventionally disclosed one.

For common 32 QAM signal, the 60-mile-radius service area can beestablished theoretically. The signal level will however be attenuatedby geographical or weather conditions and particularly, considerablydeclined at near the limit of the service area.

If the low frequency band TV component of MPEG1 grade is carried on the1-1 level D₁₋₁ data and the medium frequency band TV component of NTSCgrade on the 1-2 level D₁₋₂ data and high frequency band TV component ofHDTV on the second level D₂ data, the service area of the 32 SRQAMsignal of the present invention is increased by 10 miles in radius forreception of an EDTV signal of medium resolution grade and 18 miles forreception of an LDTV signal of low resolution grade although decreasedby 5 miles for reception of an HDTV signal of high resolution grade, asshown in FIG. 106. FIG. 107 shows a service area in case of a shiftfactor n or s=1.8. FIG. 135 shows the service area of FIG. 107 in termsof area.

More particularly, the medium resolution component of a digital TVbroadcast signal of the SRQAM mode of the preset invention cansuccessfully be intercepted in an unfavorable service region or shadowarea where a conventional medium frequency band TV signal is hardlypropagated and attenuated due to obstacles. Within at least thepredetermined service area, the NTSC TV signal of the SRQAM mode can beintercepted by any traditional TV receiver. As the shadow or signalattenuating area developed by building structures and other obstacles orby interference of a neighbor analogue TV signal or produced in a lowland is decreased to a minimum, TV viewers or subscribers will beincreased in number.

Also, the HDTV service can be appreciated by only a few viewers whoafford to have a set of high cost HDTV receiver and display, accordingto the conventional system. The system of the present invention allows atraditional NTSC, PAL, or SECAM receiver to intercept a mediumresolution component of the digital HDTV signal with the use of anadditional digital tuner. A majority of TV viewers can hence enjoy theservice at less cost and will be increased in number. This willencourage the TV broadcast business and create an extra social benefit.

Furthermore, the signal receivable area for medium resolution or NTSC TVservice according to the present invention is increased about 36% atn=2.5, as compared with the conventional system, . As the service areathus the number of TV viewers is increased, the TV broadcast businessenjoys an increasing profit. This reduces a risk in the development of anew digital TV business which will thus be encouraged to put intopractice.

FIG. 107 shows the service area of a 32 SRQAM signal of the presentinvention in which the same effect will be ensured at n=1.8. Two serviceareas 703a and 703b, of D₁ and D₂ signals respectively can be determinedin extension for optimum signal propagation by varying the shift nconsidering a profile of HDTV and NTSC receiver distribution orgeographical features. Accordingly, TV viewers will satisfy the serviceand a supplier station will enjoy a maximum of viewers.

This advantage is given when:n>1.0Hence, if the 32 SRQAM signal is selected, the shift n is determined by:1<n<5Also, if the 16 SRQAM signal is employed, n is determined by:1<n<3

In the SRQAM mode signal terrestrial broadcast service in which thefirst and second data levels are created by shifting correspondingsignal points as shown in FIGS. 99 and 100, the advantage of the presentinvention will be given when the shift n in a 16, 32, or 64 SRQAM signalis more than 1.0.

In the above embodiments, the low and high frequency band components ofa video signal are transmitted as the first and second data streams.However, the transmitted signal may be an audio signal. In this case,low frequency or low resolution components of an audio signal may betransmitted as the first data stream, and high frequency or highresolution components of the audio signal may be transmitted as thesecond data stream. Accordingly, it is possible to receive high C/Nportion in high sound quality, and low C/N portion in low sound quality.This can be utilized in PCM broadcast, radio, portable telephone and thelike. In this case, the broadcasting area or communication distance canbe expanded as compared with the conventional systems.

Furthermore, the third embodiment can incorporate a time divisionmultiplexing (TDM) system as shown in FIG. 133. Utilization of the TDMmakes it possible to increase the number of subchannels. An ECC encoder743a and ECC encoder 743b, provided in two subchannels, differentiateECC code gains so as to make a difference between thresholds of thesetwo subchannels, whereby an increase in the number of channels of themulti-level signal transmission can be realized. In this case, it isalso possible to provide the ECC encoder, such as two Trellis encoders743a and 743b for VSB-ASK signals of 4 VSB, 8 VSB, 16 VSB as shown inFIG. 137 and differentiate their code gains. The explanation of thisblock diagram is substantially identical to that of later describedblock diagram of FIG. 131 which shows the sixth embodiment of thepresent invention and, therefore, will not be described here.

FIG. 131 is a block diagram of the magnetic recording and reproducingapparatus, and FIG. 137 is a block diagram of the transmissionapparatus.

The up converter of the transmitter and the down converter of thereceiver of the transmission apparatus can be substituted for themagnetic head recording signal amplifier circuit and the magnetic headreproducing signal amplifier circuit of the magnetic recording andreproducing apparatus, respectively, and there respective components aretherefore identically constructed. The configuration and operation ofthe modulator and demodulator of the magnetic recording and reproducingapparatus are therefore also identical to those of the transmissionapparatus. Similarly, the recording/reproducing/transmission systemshown in FIG. 84 is identical in construction to the transmission systemshown in FIG. 156. To further simplify the system, the configurationshown in the block diagram of FIG. 157 can be used, or for even greatersimplification the block diagram of FIG. 158 can be used.

In a simulation of FIG. 106, there is provided 5 dB difference of acoding gain between 1-1 subchannel D₁₋₁ and 1-2 subchannel D₁₋₂.

An SRQAM is the system applying a C-CDM (Constellation-Code DivisionMultiplex) of the present invention to a rectangle-QAM. A C-CDM, whichis a multiplexing method independent of TDM or FDM, can obtainsubchannels by dividing a constellation-code corresponding to a code. Anincrease of the number of codes will bring an expansion of transmissioncapacity, which is not attained by TDM or FDM alone, while maintainingalmost perfect compatibility with conventional communication apparatus.Thus C-CDM can bring excellent effects.

Although above embodiment combines the C-CDM and the TDM, it is alsopossible to combine the C-CDM with the FDM (Frequency DivisionMultiplex) to obtain similar modulation effect of threshold values. Sucha system can be used for a TV broadcasting, and FIGS. 108(a)-108(e) showshows a frequency distribution of a TV signal. A spectrum 725 representsa frequency distribution of a conventional analogue, e.g. NTSC,broadcasting signal. The largest signal is a video carrier 722. A colorcarrier 723 and a sound carrier 724 are not so large. There is known amethod of using an FDM for dividing a digital broadcasting signal intotwo frequencies. In this case, a carrier is divided into a first carrier726 and a second carrier 727 to transmit a first 720 and a second signal721 respectively. Interference can be lowered by placing first andsecond carriers 726 and 727 sufficiently far from the video carrier 722.The first signal 720 serves to transmit a low resolution TV signal at alarge output level, while the second signal 721 serves to transmit ahigh resolution TV signal at a small output level. Consequently, themulti-level signal transmission making use of an FDM can be realizedwithout being bothered by obstruction.

FIG. 134 shows an example of a conventional method using a 32 QAMsystem. As the subchannel A has a larger output than the subchannel B, athreshold value for the subchannel A, i.e. a threshold 1, can be setsmall 4˜5 dB than a threshold value for the subchannel B, i.e. athreshold 2. Accordingly, a two-level broadcasting having 4˜5 dBthreshold difference can be realized. In this case, however, a largereduction of signal reception amount will occur if the receiving signallevel decreases below the threshold 2. Because the second signal 721a,having a large information amount as shaded in the drawing, cannot bereceived in such a case and only the first signal 720a, having a smallinformation amount, is received. Consequently, a pecturepicture qualitybrought by the second level will be extremely worse.

However, the present invention resolves this problem. According to thepresent invention, the first signal 720 is given by 32 SRQAM mode whichis obtained through C-CDM modulation so that the subchannel A is dividedinto two subchannels 1 of A and 2 of A. The newly added subchannel 1 ofA, having a lowest threshold value, carries a low resolution component.The second signal 721 is also given by 32 SRQAM mode, and a thresholdvalue for the subchannel 1 of B is equalized with the threshold 2.

With this arrangement, the region in which a transmitted signal is notreceived when the signal level decreases below the threshold 2 isreduced to a shaded portion of the second signal 721a in FIG. 108. Asthe subchannel 1 of B and the subchannel A are both receivable, thetransmission amount is not so much reduced in total. Accordingly, abetter picture quality is reproduced even in the second level at thesignal level of the threshold 2.

By transmitting a normal resolution component in one subchannel, itbecomes possible to increase the number of multiple level and expand alow resolution service area. This low-threshold subchannel is utilizedfor transmitting important information such as sound information, syncinformation, headers of respective data, because these informationcarried on this low-threshold subchannel can be surely received. Thusstable reception is feasible. If a subchannel is newly added in thesecond signal 721 in the same manner, the number of levels ofmulti-level transmission can be increased in the service area. In thecase where an HDTV signal has 1050 scanning lines, an a new service areaequivalent to 775 lines can be provided in addition to 525 lines.

Accordingly, the combination of the FDM and the C-CDM realizes anincrease of service area. Although above embodiment divides a subchannelinto two, it is needless to say it is also possible to divide it intothree or more parts.

Next, a method of avoiding obstruction by combining the TDM and theC-CDM will be explained. As shown in FIG. 109, an analogue TV signalincludes a horizontal retrace line portion 732 and a video signalportion 731. This method utilizes a low signal level of the horizontalretrace line portion 732 and non-display of obstruction on a pictureplane during this period. By synchronizing a digital TV signal with ananalogue TV signal, horizontal retrace line sync slots 733 and 733a ofthe horizontal retrace line portion 732 can be used for transmission ofan important signal, e.g. a sync signal or numerous data at a highoutput level. Thus, it becomes possible to increase the data amount oroutput level without increasing obstruction. A similar effect will beexpected even if vertical retrace line sync slots 737 and 737a areprovided synchronously with vertical retrace line portions 735 and 735a.

FIG. 110 shows a principle of the C-CDM. Furthermore, FIG. 111 shows acode assignment of the C-CDM equivalent to an expanded 16 QAM. FIG. 112shows a code assignment of the C-CDM equivalent to an expanded 32 QAM.As shown in FIGS. 110 and 111, a 256 QAM signal is divided into four,740a, 740b, 740c, and 740d, levels which have 4, 16, 64, and 256segments, respectively. A signal code word 742d of 256 QAM on the fourthlevel 740d is “11111111” of 8 bits. This is split into four code words741a, 741b, 741c, and 741d of 2-bits—i.e. “11”, “11”, “11”, “11”, whichare then allocated on signal point regions 742a, 742b, 742c, and 742d offirst, second, third, and fourth levels 740a, 740b, 740c, and 740d,respectively. As a result, subchannels 1, 2, 3, and 4 of 2 bits arecreated. This is termed C-CDM (Constellation-Code Division Multiplex).FIG. 111 shows a detailed code assignment of the C-CDM equivalent toexpanded 16 QAM, and FIG. 112 shows a detailed code assignment of theC-CDM equivalent to expanded 32 QAM. As the C-CDM is an independentmultiplexing method, it can be combined with the conventional FDM(Frequency Division Multiplex) or TDM (Time Division Multiplex) tofurther increase the number of subchannels. In this manner, the C-CDMmethod realizes a novel multiplexing system. Although the C-CDM isexplained by using rectangular QAM, other modulation system havingsignal points, e.g. QAM, PSK, ASK, and even FSK if frequency regions areregarded as signal points, can be also used for this multiplexing in thesame manner.

For example, the error rate of the subchannel 1 of 8PS-APSK, explainedin the embodiment 1 with reference to FIG. 139, will be expressed asfollows:${\text{[}{Pe}_{1 - 8}} = {{\frac{1}{4}{{erfc}\left( \frac{\delta}{\sqrt{2\alpha}} \right)}} + {\frac{1}{4}{{erfc}\left( \frac{\left( {S_{1} + 1} \right)\delta}{\sqrt{2\sigma}} \right)}\text{]}}}$${Pe}_{1 - 8} = {{\frac{1}{4}{{erfc}\left( \frac{\delta}{\sqrt{2\sigma}} \right)}} + {\frac{1}{4}{{erfc}\left( \frac{\left( {S_{1} + i} \right)\delta}{\sqrt{2\sigma}} \right)}}}$

The error rate of the subchannel 2 is expressed as follows:${Pe}_{2 - 8} = {\frac{1}{2}{{erfc}\left( \frac{S_{1}\delta}{2\sigma} \right)}}$

Furthermore, the error rate of the subchannel 1 of 16-PS-APSK (PS type),explained with reference to FIG. 142, will be expressed as followfollows:${Pe}_{1 - 16} = {{\frac{1}{8}{{erfc}\left( \frac{\delta}{\sqrt{2\sigma}} \right)}} + {\frac{1}{8}{{erfc}\left( \frac{\left( {S_{2} + 1} \right)\delta}{\sqrt{2\sigma}} \right)}} + {\frac{1}{8}{{erfc}\left( \frac{\left( {S_{1} + 1} \right)\delta}{\sqrt{2\sigma}} \right)}} + {\frac{1}{8}{{erfc}\left( \frac{\left( {S_{1} + S_{2} + 1} \right)\delta}{\sqrt{2\sigma}} \right)}}}$

The error rate of the subchannel 2 is expressed as follows:${Pe}_{2 - 16} = {{\frac{1}{4}{{erfc}\left( \frac{S_{1}\delta}{2\sigma} \right)}} + {\frac{1}{8}{{erfc}\left( \frac{\left( {S_{1} - S_{2}} \right)\delta}{2\sigma} \right)}} + {\frac{1}{8}{{erfc}\left( \frac{\left( {S_{1} + S_{2}} \right)\delta}{2\sigma} \right)}}}$The error rate of the subchannel 3 is expressed as follows:${Pe}_{3 - 10} = {\frac{1}{2}{{erfc}\left( \frac{S_{2}\delta}{2\sigma} \right)}}$

EMBODIMENT 4

A fourth embodiment of the present invention will be described referringto the relevant drawings.

FIG. 37 illustrates the entire arrangement of a signal transmissionsystem of the fourth embodiment, which is arranged for terrestrialservice and similar in both construction and action to that of the thirdembodiment shown in FIG. 29. The difference is that the transmitterantenna 6 is replaced by a terrestrial antenna 6a and the receiverantennas 22, 23, and 24 32, and 42are also replaced by three terrestrialantennas 22a, 23a, and 24a32a, and 42a. The action of the system isidentical to that of the third embodiment and will not be explained inmore detail. The terrestrial broadcast service unlike a satelliteservice depends much on the distance between the transmitter antenna 6ato the receiver antennas 22a, 32a, and 42a. If a receiver is located farfrom the transmitter, the level of a received signal is low.Particularly, a common multi-level QAM signal can hardly be demodulatedby the receiver which thus reproduces no TV program.

The signal transmission system of the present invention allows the firstreceiver 23 equipped with the antenna 22a, which is located at a fardistance as shown in FIG. 37, to intercept a modified 16 or 64 QAMsignal and demodulate at 4 PSK mode the first data stream or D₁component of the received signal to an NTSC video signal so that a TVprogram picture of medium resolution can be displayed even if the levelof the received signal is relatively low.

Also, the second receiver 33 with the antenna 32a is located at a mediumdistance from the antenna 6a and can thus intercept and demodulate bothfirst and second data streams or D₁ and D₂ components of the modified 16or 64 QAM signal to an HDTV video signal which in turn produces an HDTVprogram picture.

The third receiver 43 with the antenna 42a is located at a near distanceand can intercept and demodulate the first, second, and third datastreams or D₁, D₂, and D₃ components of the modified 16 or 64 QAM signalto a super HDTV video signal which in turn produces a super HDTV picturein quality to a common movie picture.

The assignment of frequencies is determined by the same manner as of thetime division multiplexing shown in FIGS. 34, 35, and 36. Like FIG. 34,when the frequencies are assigned to first to sixth channels, L1 of theD₁ component carries an NTSC data of the first channel, M1 of the D₂component carries an HDTV difference data of the first channel, and H1of the D₃ component carries a super HDTV difference data of the firstchannel. Accordingly, NTSC, HDTV, and super HDTV data all can be carriedon the same channel. If D₂ and D₃ of the other channels are utilized asshown in FIGS. 35 and 36, more data of HDTV and super HDTV respectivelycan be transmitted for higher resolution display.

As understood, the system allows three different but compatible digitalTV signals to be carried on a single channel or using D₂ and D₃ regionsof other channels. Also, the medium resolution TV picture data of eachchannel can be intercepted in a wider service area according to thepresent invention.

A variety of terrestrial digital TV broadcast systems employing a 16 QAMHDTV signal of 6 MHz bandwidth have been proposed. Those are however notcompatible with the existing NTSC system and thus, have to be associatedwith a simulcast technique for transmitting NTSC signals of the sameprogram on another channel. Also, such a common 16 QAM signal limits aservice area. The terrestrial service system of the present inventionallows a receiver located at a relatively far distance to interceptsuccessfully a medium resolution TV signal with no use of an additionaldevice nor an extra channel.

FIG. 52 shows an interference region of the service area 702 of aconventional terrestrial digital HDTV broadcast station 701. As shown,the service area 702 of the conventional HDTV station 701 intersectswith the service area 712 of a neighboring analogue TV station 711. Atthe intersecting region 713, an HDTV signal is attenuated by signalinterference from the analogue TV station 711 and will thus beintercepted with less consistency.

FIG. 53 shows an interference region associated with the multi-levelsignal transmission system of the present invention. The system is lowin the energy utilization as compared with a conventional system and itsservice area 703 for HDTV signal propagation is smaller than the area702 of the conventional system. On the contrary, the service area 704for digital NTSC or medium resolution TV signal propagation is largerthan the conventional area 702. The level of signal interference from adigital TV station 701 of the system to a neighboring analogue TVstation 711 is equivalent to that from a conventional digital TVstation, such as shown in FIG. 52.

In the service area of the digital TV station 701, there are threeinterference regions developed by signal interference from the analogueTV station 711. Both HDTV and NTSC signals can hardly be intercepted inthe first region 705. Although fairly interfered, an NTSC signal may beintercepted at an equal level in the second region 706 denoted by theleft down hatching. The NTSC signal is carried on the first data streamwhich can be reproduced at a relatively low C/N rate and will thus beminimum minimally affected when the C/N rate is declined by signalinterference from the analogue TV station 711.

At the third region 707 denoted by the right down hatching, an HDTVsignal can also be intercepted when signal interference is absent whilethe NTSC signal can constantly be intercepted at a low level.

Accordingly, the overall signal receivable area of the system will beincreased although the service area of HDTV signals becomes a little bitsmaller than that of the conventional system. Also, at the signalattenuating regions produced by interference from a neighboring analogueTV station, NTSC level signals of an HDTV program can successfully beintercepted as compared with the conventional system where no HDTVprogram is viewed in the same area. The system of the present inventionmuch reduces the size of signal attenuating area and when increases theenergy of signal transmission at a transmitter or transponder station,can extend the HDTV signal service area to an equal size to theconventional system. Also, NTSC level signals of a TV program can beintercepted more or less in a far distance area where no service isgiven by the conventional system or a signal interference area caused byan adjacent analogue TV station.

Although the embodiment employs a two-level signal transmission method,a three-level method such as shown in FIG. 78 will be used with equalsuccess. If an HDTV signal is divided into three picture levels-HDTV,NTC, and low resolution NTSC, the service area shown in FIG. 53 will beincreased from two levels to three levels where the signal propagationis extended radially and outwardly. Also, low resolution NTSC signalscan be received at an acceptable level at the first signal interferenceregion 705 where NTSC signals are hardly be intercepted in the two-levelsystem. As understood, the signal interference is also involved from adigital TV station to an analogue TV station.

The description will now be continued, provided that no digital TVstation should cause a signal interference to any neighboring analogueTV station. According to a novel system under consideration in U.S.A.,no-use channels of the existing service channels are utilized for HDTVand thus, digital signals must not interfere with analogue signals. Forthis purpose, the transmitting level of a digital signal has to bedecreased lower than that shown in FIG. 53. If the digital signal is ofconventional 16 QAM or 4 PSK mode, its HDTV service area 708 becomesdecreased as the signal interference region 713 is fairly large as shownin FIG. 54. This results in a less number of viewers and sponsors,whereby such a digital system will have much difficulty in operating asa profitable business.

FIG. 55 shows a similar result according to the system of the presentinvention. As apparent, the HDTV signal receivable area 703 is a littlebit smaller than the equal area 708 of the conventional system. However,the lower resolution or NTSC TV signal receivable area 704 will beincreased as compared with the conventional system. The hatching arearepresents a region where the NTSC level signal of a program can bereceived while HDTV signal of the semesame is hardly intercepted. At thefirst interference region 705, both HDTV and NTSC signals cannot beintercepted due to signal interference from an analogue station 711.

When the level of signals is equal, the multi-level transmission systemof the present invention provides a smaller HDTV service area and agreater NTSC service area for interception of an HDTV program at an NTSCsignal level. Accordingly, the overall service area of each station isincreased and more viewers can enjoy its TV broadcasting service.Furthermore, HDTV/NTSC compatible TV business can be operated witheconomical advantages and consistency. It is also intended that thelevel of a transmitting signal is increased when the control on avertingsignal interference to neighboring analogue TV stations is lessenedcorresponding to a sharp increase in the number of home-use digitalreceivers. Hence, the service area of HDTV signals will be increased andin this respect, the two different regions for interception of HDTV/NTSCand NTSC digital TV signal levels respectively, shown in FIG. 55, can beadjusted in proportion by varying the signal point distance in the firstand/or second data stream. As the first data stream carries informationabout the signal point distance, a multi-level signal can be receivedwith more certainty.

FIG. 56 illustrates signal interference between two digital TV stationsin which a neighboring TV station 701a also provides a digital TVbroadcast service, as compared with an analogue station in FIG. 52.Since the level of a transmitting signal becomes high, the HDTV serviceor high resolution TV signal receivable area 703inis increased to anextension equal to the service area 702 of an analogue TV system.

At the intersecting region 714 between two service areas of theirrespective stations, the received signal can be reproduced not to anHDTV level picture using a common directional antenna due to signalinterference but to an NTSC level picture with a particular directionalantenna directed towards a desired TV station. If a highly directionalantenna is used, the received signal from a target station will bereproduced as an HDTV picture. The low resolution signal receivable area704 is increased larger than the analogue TV system service area 702 anda pair of intersecting regions 715 and 716 developed by the two lowresolution signal receivable areas 704 and 704a of their respectivedigital TV stations 701 and 701a permit the received signal from anantenna directed to one of the two stations to be reproduced as an NTSClevel picture.

The HDTV service area of the multi-level signal transmission system ofthe present invention itself will be much increased when applicablesignal restriction rules are withdrawn in a coming digital TV broadcastservice maturity time.

At the time, the system of the present invention also provides as a wideHDTV signal receivable area as of the conventional system andparticularly, allows its transmitting signal to be reproduced at an NTSClevel in a further distance or intersecting areas where TV signals ofthe conventional system are hardly intercepted. Accordingly, signalattenuating or shadow regions in the service area will be minimized.

EMBODIMENT 5

A fifth embodiment of the present invention resides in amplitudemodulation or ASK procedure. FIG. 57 illustrates the assignment ofsignal points of a 4-level ASK signal, such as VSB signal, according tothe fifth embodiment, in which four signal points are denoted by 721,722, 723, and 724. FIG. 68(a) shows the constellation of 8-level VSBsignal. The four-level transmission permits a 2-bit data to betransmitted in every cycle period where as the eight-level transmissionpermits a 4-bit data. It is assumed that the four signal points 721,722, 723, and 724 in the case of 4 VSB represent two-bit patterns 00,01, 10, and 11 respectively.

In FIG. 58, the constellation of 4-level ASK, such as 4-level VSB, isshown. For ease of four-level signal transmission of the embodiment, thetwo signal points 721 and 722 are designated as a first signal pointgroup 725 and the other two signal points 723 and 724 are designated asa second signal point group 726. The distance between the two signalpoint groups 725 and 726 is then determined wider than that between anytwo adjacent signal points. More specifically, the distance L₀ betweenthe two signals 722 and 723 is arranged wider than the distance Lbetween the two adjacent points 721 and 722 or 723 and 724.

This is expressed as:L₀>L

Hence, the multi-level signal transmission system of the embodiment isbased on L₀>L. The embodiment is however not limited to L₀>L, and L₀=Lwill be employed temporarily or permanently depending on therequirements of design, condition, and setting. In the case of VSB, theconstellation shown in FIGS. 68(a) and (b) are taken.

The two signal point group groups are assigned one-bit patterns of thefirst data stream D₁, as shown in FIG. 59(a). More particularly, a bit 0of binary system is assigned to the first signal point group 725 andanother bit 1 to the second signal point group 726. Then, a one-bitpattern of the second data stream D₂ is assigned to each signal point.For example, the two signal points 721 and 723 are assigned D₂=0 and theother two signal points 722 and 724 are assigned D₂=1.

The multi-level signal transmission of the present invention can beimplemented in an ASK mode using the foregoing signal point assignment.The system of the present invention works in the same manner as of aconventional equal signal point distance technique when the signal tonoise ratio or C/N rate is high. If the C/N rate becomes low and no datacan be reproduced by the conventional technique, the present systemensures reproduction of the first data stream D₁ but not the second datastream D₂. In more detail, the state at a low C/N is shown in FIG. 60illustrating the constellation of ASK of 4 VSB. The signal pointstransmitted are displaced by a Gaussian distrigution distribution toranges 721a, 722a, 723a, and 724a respectively at the receiver side dueto noise and transmission distortion. Therefore, the distinction betweenthe two signals 721 and 722 in the case of slice level 2 or between 723and 724 in the case of slice level 4 will hardly be executed. In otherwords, the error rate in the second data stream D₂ will be increased. Asapparent from FIG. 60, the two signal points 721 and 722 are easilydistinguished from the other two signal points 723 and 724. Thedistinction between the two signal point groups 725 and 726 can thus becarried out with ease. As the result, the first data stream D₁ will bereproduced at a low error rate.

Accordingly, the two different level data D₁ and D₂ can be transmittedsimultaneously. More particularly, both the first and second datastreams D₁ and D₂ of a given signal transmitted through the multi-leveltransmission system can be reproduced at the area where the C/N rate ishigh and the first data stream D₁ only can be reproduced in the areawhere the C/N rate is low.

FIG. 61 is a block diagram of a transmitter 741 in which an input unit742 comprises a first data stream input 743 and a second data streaminput 744. A carrier wave from a carrier generator 64 is amplitudemodulated by a multiplier 746 using an input signal fed through aprocessor 745 from the input unit 743 742to provide the 4-level or8-level ASK signal, as shown in FIG. 62(a). The modulated signal, i.e.,the 4-level or 8-level ASK signal is then band limited by a band passfilter 747 into a vestigial side band having some residual side band ofthe carrier, as shown in FIG. 62(b), i.e., to an ASK signal of e.g. VSBmode which is then delivered from an output unit 748.

The waveform of the ASK signal after filtering will now be examined.FIG. 62(a) shows a frequency spectrum of the ASK modulated signal inwhich two sidebands are provided on both sides of the carrier frequencyband. One of the two sidebands is eliminated by the filter 474 toproduce a signal 749 which contains a carrier component as shown in FIG.62(b). The signal 749 is a VSB signal and if the modulation frequencyband is f₀, will be transmitted in a frequency band of about f₀/2.Hence, the frequency utilization becomes high. Using VSB modetransmission, the ASK signal of two bit per symbol shown in FIG. 60 canthus carry in the same frequency band an amount of data equal to that of16 QAM mode at four bits per symbol for 4 VSB, and 32 QAM mode at fivebits per symbol for 8 VSB.

FIG. 63 is a block diagram of a receiver 751 in which an input signalintercepted by a terrestrial antenna 32a is transferred through an inputunit 752 to a mixer 753 where it is mixed with a signal from a variableoscillator 754 controlled by channel selection to a lower mediumfrequency signal. The signal from the mixer 753 is then detected by adetector 755 and filtered by an LPF 756 to a baseband signal which istransferred to a discriminating/demodulator circuit 757 which has4-level slicer in the case of 4 VSB, and 8-level slicer in the case of 8VSB. The discrimination/demodulator circuit 757 reproduces two, first D₁and second D₂, data streams from the baseband signal and transmits themfurther through first and second data stream output 758 and 759respectively.

The transmission of a TV signal using such a transmitter and a receiverwill be explained. FIG. 64 is a block diagram of a video signaltransmitter 774 in which a high resolution TV signal, e.g. an HDTVsignal, is fed through an input unit 403 to a divider circuit 404 of afirst video encoder 401 where it is divided into four high/low frequencyTV signal components denoted by e.g. H_(L)V_(L), H_(L)V_(H), H_(H)V_(L),and H_(H)V_(H). This action is identical to that of the third embodimentpreviously described referring to FIG. 30 and will not be explained inmore detail. The four separate TV signals are encoded respectively by acompressor 405 using a known DPCMDCT variable length code encodingtechnique which is commonly used e.g. in MPEG. Meanwhile, the motioncompensation of the signal is carried out at the input unit 403. Thecompressed signals are summed by a summer 771 into two, first andsecond, data streams D₁ and D₂. The low frequency video signal componentor H_(L)V_(L) signal is contained in the first data stream D₁. The twodata stream signals D₁ and D₂ are then transferred to first and seconddata stream inputs 743 and 744 of a transmitter unit 741 where they areamplitude modulated and summed into an ASK signal of e.g. VSB mode whichis propagated from a terrestrial antenna for broadcast service.

FIG. 65 is a block diagram of a TV receiver for such a digital TVbroadcast system. A 4-VSB or 8-VSB digital TV signal intercepted by aterrestrial antenna 32a is fed to an input 752 of a receiver 781. Thesignal is then transferred to a VSB detection/demodulation circuit 760where a desired channel signal is selected and demodulated to two, firstand second, data streams D₁ and D₂ which are then fed to first andsecond data stream outputs 758 and 759 respectively. The operation inthe receiver unit 751 is similar to that described previously and willnot be explained in more detail. The two data streams D₁ and D₂ are sentto a divider unit 776 in which D₁ is divided by a divider 777 into twocomponents; one or compressed H_(L)V_(L) is transferred to a first input521 of a second video decoder 422 and the other is fed to a summer 778where it is summed with D₂ prior to transfer to a second input 531 ofthe second video decoder 422. Compressed H_(L)V_(L) is then sent fromthe first input 521 to a first expander 523 where it is expanded toH_(L)V_(L) of the original length which is then transferred to a videomixer 548 and an aspect ratio changing circuit 779. When the input TVsignal is an HDTV signal, H_(L)V_(L) represents a wide-screen NTSCsignal. When the same is an NTSC signal, H_(L)V_(L) represents a lowerresolution video signal, e.g. MPEG1, that than an NTSC level.

The input TV signal of the embodiment is an HDTV signal and H_(L)V_(L)becomes a wide-screen NTSC signal. If the aspect ratio of an availabledisplay is 16:9, H_(L)V_(L) is directly delivered through an output unitas a 16:9 video output 426. If the display has an aspect ratio of 4:3,H_(L)V_(L) is shifted by the aspect ratio changing circuit 779 to aletterbox or sidepanel format and then, delivered from the output unit780 as a corresponding format video output 425.

The second data stream D₂ fed from the second data stream output 759 tothe summer 778 is summed with the output of the divider 777 to a sumsignal which is then fed to the second input 531 of the second videodecoder 422. The sum signal is further transferred to a divider circuit531 while it is divided into three compressed forms of H_(L)V_(H),H_(H)V_(L), and H_(H)V_(H). The three compressed signals are then fed toa second 535, a third 536, and a fourth expander 537 respectively forconverting by expansion to H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H), ofthe original length. The three signals are summed with H_(L)V_(L), bythe video mixer 548 to a composite HDTV signal which is fed through anoutput 546 of the second video decoder to the output unit 780. Finally,the HDTV signal is delivered from the output unit 780 as an HDTV videosignal 427.

The output unit 780 is arranged for detecting an error rate in thesecond data stream of the second data stream output 759 through an errorrate detector 782 and if a condition in which the error rate is highcontinues for a predetermined time, H_(L)V_(L) of low resolution videodata systematically are produced for a predetermined time.

Accordingly, the multi-level signal transmission system for digital TVsignal transmission and reception becomes feasible. For example, if a TVsignal transmitter station is near, both the first and second datastreams of a received signal can successfully be reproduced to exhibitan HDTV quality picture. If the transmitter station is far, the firstdata stream can be reproduced to H_(L)V_(L) which is converted to a lowresolution TV picture. Hence, any TV program will be intercepted in awider area and displayed at a picture quality ranging from HDTV to NTSClevel.

FIG. 66 is a block diagram showing another arrangement of the TVreceiver. As shown, the receiver unit 751 contains only a first datastream output 768 and thus, the processing of the second data stream orHDTV data is not needed so that the overall construction can beminimized. It is a good idea to have the first video decoder 421 shownin FIG. 31 as a video decoder of the receiver. Accordingly, an NTSClevel picture will be reproduced. The receiver is fabricated at muchless cost as having no capability to receive any HDTV level signal andwill widely be accepted in the market. In brief, the receiver can beused as an adapter tuner for interception of a digital TV signal withgiving no modification to the existing TV system including a display.

When a scrambled 4-level VSB or 8-level VSB is received as shown in FIG.66, the scramble cancellation signal transmitted with the 4- or 8-levelVSB signal is compared by the descramble number comparator 502b with thenumber stored in the descramble number register 502c in the descrambler502. Only when the transmitted stored numbers match is descrambling ofspecific scrambled transmissions permitted.

The TV receiver 781 may have a further arrangement shown in FIG. 67,which serves as both a satellite broadcast receiver for demodulation ofPSK signals and a terrestrial broadcast receiver for demodulation of VSBsignals. In operation, a PSK signal received by a satellite antenna 32is mixed by a mixer 786 with a signal from an oscillator 787 into a lowfrequency signal which is then fed through an input unit 34 to a mixer735 similar to one shown in FIG. 63. The low frequency signal of PSK orQAM mode in a given channel of the satellite TV system is transferred toa modulator demodulator 35 where two data streams D₁ and D₂ arereproduced from the signal. D₁ and D₂ are sent through a divider 788 toa second video decoder 422 where they are converted to a video signalwhich is then delivered from an output unit 780. Also, a digital oranalogue terrestrial TV signal intercepted by a terrestrial antenna 32ais fed through an input unit 752 to the mixer 753 where one desiredchannel is selected in the same manner as described in FIG. 63 andconverted into to a low frequency base band signal. The signal ofanalogue form is sent directly to the demodulator 35 for demodulation.The signal of digital form is then fed to a discrimination/demodulationcircuit 757 where two data streams D₁ and D₂ are reproduced from thesignal. D₁ and D₂ are converted by the second video decoder 422 into avideo signal which is then delivered further. A satellite analogue TVsignal is transferred to a video demodulator 788 7880 where it is AN AMmodulated into an analogue video signal which is then delivered from theoutput unit 780. As understood, the mixer 753 of the TV receiver 781shown in FIG. 67 is arranged to be compatible between two, satellite andterrestrial, broadcast services. Also, a receiver circuit including adetector 755 and an LPF 756 for AM modulation of an analogue signal canbe utilized compatible with a digital ASK signal of the terrestrial TVserivice service. The major part of the arrangement shown in FIG. 67 isarranged for compatible use, thus minimizing a circuitry construction.

According to the embodiment, a 4-level ASK signal is divided into two,D₁ and D₂, level components for execution of the one-bit modemulti-level signal transmission. If an 8-level ASk ASK signal as shownin FIGS. 68(a) and (b) illustrating the constellation of the 8-VSBsignal, i.e., 8-level VSB signal is used, it can be transmitted in aone-bit mode three-level, D₁, D₂, and D₃, arrangement, thus, three bitspre per symbol in total. A shown in FIG. 68(a), the first bit coding isdone as follows. D₃ is assigned to eight signal points 721a and 721b;722a and 722b; 723a and 723b; and 724a and 724b, each pair, i.e., smallgroup, representing a two-level pattern using one bit. Next, the secondbit coding is done as follows. D₂ is assigned to two signal point groups721 and 722; and 723 and 724, two mid groups representing a two-levelpattern using one bit. Next, the third bit coding is done as follows.D₁, is assigned to two large signal point groups 725 and 726representing a two-level pattern using one bit. More particularly, thisis equivalent to a form in which each of the four signal points 721,722, 723, and 724 shown in FIG. 57 is divided into two components thusproducint producing, at the maximum, three different level data.

As understood from the above, each of the eight signal points isassigned with three bit data (D₁, D₂, D₃). For example, if left side andright side are defined as logic 0 and logic 1, respectively, the threebit data (D₁, D₂, D₃) for the signal point 722a will be (0, 1, 0). Thiscan be explained as follows. Since the signal point 722a is in the leftside of the two large groups 725 and 726, logic 0 is given to D₁. Also,the signal point 722a is in the right side of the two mid groups 721 and722, so that logic 1 is given to D₂. Further the signal point 722a is inthe left side of the two small groups 722a and 722b, so that logic 0 isgiven to D₃. In a similar manner, the three bit data (D₁, D₂, D₃) forthe signal point 723a will be (1, 0, 0).

The three-level signal transmission, such as for the digital HDTV, isidentical to that described in the third and fourth embodiments and willno further be explained in detail.

The effects of television broadcasting using the 8-level VSB shown inFIGS. 68(a), (b), and (c) are described below.

While the transmitted data quantity is high with 8-level VSB, it alsohas a higher error rate than 4-level VSB for the same C/N value.However, in high image quality HDTV transmissions, the availabletransmission capacity makes it possible to apply more error correctioncoding, and the error rate can thus be reduced. This band capacity alsoenable enables multi-level (hierarchical) television broadcasts andother new features in the future.

The relative effects of 4-level, 8-level, and 16-level VSB are describedbelow.

In ground station broadcasts using the NTSC or PAL frequency band, theusable transmission band is effectively limited to approximately 5 MHzbecause of the 6-MHz frequency limit of the NTSC format, for example, asshown in FIG. 136. With 4-level VSB, the effective data transmissionquantity is 5 MHz×4=20 Mbps because the frequency utilization efficiencyis 4 bits/Hz. A minimum of 15 Mbps-18 Mbps is required, however, fordigital HDTV signal transmission. Because there is no spare capacitywith 4-level VSB, the redundancy used for error correction is only10-20% of the HDTV effective transmission quantity as shown in thecomparison chart in FIG. 169.

With 8-level VSB, the effective data transmission quantity is 5 MHz×6=30Mbps because the frequency utilization efficiency is 5 bits/Hz. While 15Mbps-18 Mbps is required for digital HDTV signal transmission asdescribed above, when using 8-level VSB modulation, more than 50% of theactual HDTV signal transmission quantity can be used for errorcorrection coding as shown in FIG. 169. As shown by error rate curves805 and 806 in FIG. 161 163, the error rate relative to the same C/Nvalue in the transmission system is less with TCM 8-level VSB than with4-level VSB, even through error correction code gain is greater with8-level VSB than with 4-level VSB, because significantly more errorcorrection coding can be added with 8-level VSB during ground stationbroadcasting of same-data-rate HDTV digital signals using the 6-MHzband. As a result, 8-level VSB with high code gain error correctioncoding also has the effect of enabling a larger service area for groundstation HDTV broadcasts than does 4-level VSB. While the increased sizeof the error correction circuits required with 8-level VSB does increasethe complexity of the receiver circuitry, the circuit scale of theequalizer in the receiver is significantly smaller than that ofreceivers using QAM modulation, which contains a phase component,because VSB and ASK are amplitude modulation methods. As a result, an8-level VSB circuit board containing the error correction circuit issmaller than an equivalent 32-level QAM board with the same transmissioncapacity.

A digital HDTV receiver with an appropriate circuit scale and a largeground station broadcasting service area can therefore be achieved using8-level VSB.

Note that the ECC 744a and trellis encoder 744b in the transmitter andreceiver block diagrams of FIG. 84 for the present embodiment, FIGS.131, 137, 156, and 157 for embodiment 6, and FIG. 144 for embodiment 9are used as examples of the specific error correction method, and the4-, 8-, and 16-level VSB modulator 749 described with reference to FIG.61 are used for transmission. The VSB demodulator 760 described withreference to FIG. 63 is used in the receiver to generate the digitalreception data by means of the 4-, 8-, and 16-level level slicer 757from the 4-, 8-, and 16-level VSB signal. After error correction bymeans of the trellis decoder 759b and ECC decoder 759a, described belowwith reference to FIG. 84 for the present embodiment, and FIGS. 131,137, 156, and 157 for embodiment 6, a digital HDTV signal is generatedby the image expander of the image decoder 402, and the digital HDTVsignal is then output.

As shown in FIGS. 160(a) and (b) described below with the sixthembodiment, the ECC encoder 744a uses a Reed-Solomon encoder 744j andinterleaver 744k, and uses a deinterleaver 759k and Reed-Solomon decoder759j for the ECC decoder 759a. Applying interleaving as described in theprevious embodiment improves resistance to transmission system noisesuch as burst error.

Code gain can be further increased and the error rate decreased by usinga trellis encoder as shown in FIGS. 128(a), (b), (c), (d), (e), and (f).A ratio 2/3 trellis encoder 744b 743c and decoder 759b as shown in FIG.172 are most appropriate with 8-level VSB because of 3 bits/symbolcoding. The data quantity is compressed 2/3 in this case.

The embodiments have been described using primarily the example of amultilevel (hierarchical) digital television signal. While an idealbroadcasting format can be achieved using a multilevel signal, the imagecompression circuit and modulator/demodulator circuits become morecomplex, and are therefore not preferable due to cost for the start ofthe new broadcasting services. As described at the beginning of thefifth embodiment, a broadcasting system with a simple television circuitcan be achieved by using a signal-signal interval L=L₀, i.e., an equalinterval, in the 4-level VSB and 8-level VSB signals and anon-multilevel television transmission, and by simplifying the circuitshown in FIG. 137 as shown in FIG. 157. Once the HDTV format is incommon use, it is then possible to change to a hierarchical 8-level VSBtransmission format.

Four- and 8-level VSB have been described above, and 16- and 32-levelVSB are described below with reference to FIGS. 159(a)-(b). FIG. 159(a)shows the 16-level VSB constellation. As shown in FIG. 159(b), thesignal between two signal points is grouped into eight groups 722a-722h,which are treated as eight signal points and can be treated as 8-levelVSB signals to enable a two-stage multilevel transmission. In this case,multilevel transmission can be achieved with time division multiplexingeven when intermittently transmitting an 8-level VSB signal. The maximumdata rate with this method is 2/3. In FIG. 157(c), the data is furthergrouped into four groups 723a-723d, which can be handled as 4-level VSBsignals adding one more level to the hierarchy. While the maximum datarate drops with time division multiplex transmission of 4-level VSBsignals, multilevel transmission is possible with 3-stage multilevel VSBtransmission.

With this method, a multilevel transmission whereby 8-level VSB or4-level VSB data can be reproduced when the C/N ratio of the 16-levelVSB data deteriorates can be achieved. By doubling the signal points ofthe 16-level VSB format as shown in FIG. 159(d), 32-level VSBtransmission is enabled. When 16-level VSB capacity is increased in thefuture, this method will maintain compatibility while making it possibleto obtain a 6-bit/symbol data capacity.

By summarizing the above, the VSB receiver shown in the block diagram ofFIG. 161 and the VSB transmitter shown in the block diagram of FIG. 162can be achieved.

While 4-level VSB and 8-level VSB are used by way of example above,16-level VSB as shown in FIGS. 159(a)-(c) can also be used fortransmission. With 16-level VSB, a 40-Mbps transmission capacity can beused with a 6-MHz band in ground station broadcasting. Because the datarate of the HDTV digital compression signal is 15-18 Mbps using the MPEGstandard, there is excessive reserve in the transmission capacity. Asshown in FIG. 169, redundancy R₁₆=100% or greater; redundancy istherefore too great for transmitting one channel digital HDTV, and thecircuitry is simply made more complex with little additional advantagegained over 8-level VSB. In addition, 16-level VSB redundancy is onlyabout 10%, the same as 4-level VSB redundancy, in ground station HDTVbroadcasting of two programs with 16-level VSB. As a result, the servicearea is reduced because sufficient error correction coding cannot beapplied with two-program 16-level VSB. As described before, sufficienterror correction cannot be applied with 4-level VSB because theredundancy R₄=10-20% and the service area is limited. As will be knownfrom FIG. 169, sufficient error correction coding can be achieved with8-level VSB because the redundancy R₈=50%. A broad service area can alsobe achieved without significantly .

In particular, the arrangement of the video encoder 401 of the thirdembodiment shown in FIG. 30 is replaced by a modification of the blockdiagram of FIG. 69. The operation of the modified arrangement is similarand will not be explained in greater detail. Two video signal dividercircuits 404 and 404a which may be sub-band filters are provided forminga divider unit 794. The divider unit 794 may also be arranged moresimple a simply as shown in the block diagram of FIG. 70, in which asignal passes across one signal divider circuit two times at timedivision mode. More specifically, a video signal of e.g. HDTV or superHDTV from the input unit 403 is time-base compressed by a time-basecompressor 795 and fed to the divider circuit 404 where it is dividedinto four components, H_(H)V_(H)-H, H_(H)V_(L)-H, and H_(L)V_(H)-H, andH_(L)V_(L)-H, at a first cycle. At the time, four switches 765, 765a,765b, and 765c remain turned to the position 1 so that H_(H)V_(H)-H,H_(H)V_(L)-H, and H_(L)V_(H)-H, are transmitted to a compressing circuit405. Meanwhile, H_(L)V_(L)-H is fed back through the terminal 1 of theswitch 765c to the time-base compressor 795. At a second cycle, the fourswitches 765, 765a, 765b, and 765c turned to the position 2 and all thefour components of the divider circuit 404 are simultaneouslytransferred to the compressing circuit 405. Accordingly, the dividerunit 769 794 of FIG. 70 arranged for time division processing of aninput signal can be constructed in a simpler dividing circuit form.

At the receiver side, such a video decoder as described in the thirdembodiment and shown in FIG. 30 is needed for three-level transmissionof a video signal. More particularly, a third video decoder 423 isprovided which contains two mixers 556 and 556a of different processingcapability as shown in the block diagram of FIG. 71.

Also, the third video decoder 423 may be modified in which the sameaction is executed with one single mixer 556 as shown in FIG. 72. At thefirst timing, five swatches switches 765, 765a, 765b, 765c, 765d remainsturned to the position 1. Hence, H_(L)V_(L), H_(L)V_(H), H_(H)V_(L), andH_(H)V_(H) are fed from a first 522, a second 522a, a third 522b and afourth expander 522c to through their respective switches to the mixer556 where they are mixed to a single video signal. The video signalwhich represents H_(L)V_(L)-H of an input high resolution video signalis then fed back through the terminal 1 of the switch 765d to theterminal 2 of the switch 765c. At the second timing, the four switches765, 765a, 765b, 765c are turned to the point 2. Thus, H_(H)V_(H)-H,H_(H)V_(L)-H, H_(L)V_(H)-H, and H_(L)V_(L)-H are transferred to themixer 556 where they are mixed to a single video signal which is thensent across the terminal 2 of the switch 765d to the output unit 554 forfurther delivery.

In this manner of time division processing of a three-level signal, twomixers can be replaced with one mixer.

More particularly, four components H_(L)V_(L), H_(L)V_(H), H_(H)V_(L),and H_(H)V_(H) are fed to produce H_(L)V_(L)-H at the first timing.Then, H_(L)V_(H)-H, H_(H)V_(L)-H, and H_(H)V_(H)-H are fed at the secondtiming delayed from the first timing and mixed with H_(L)V_(L)-H to atarget video signal. It is thus essential to perform the two actions atan interval of time.

If the four components are overlap each other or are supplied in avariable sequence, they have to be time-base adjusted to a givensequence through using memories accompanied with their respectiveswitches 765, 765a, 765b, and 765c. In the foregoing manner, a signal istransmitted from the transmitter at two different timing periods asshown in FIG. 73 so that no time-base controlling circuit is needed inthe receiver which is thus arranged more compact compactly.

As shown in FIG. 73, D₁ is the first data stream of a transmittingsignal and H_(L)V_(L), H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) aretransmitted on D₁ channel at the period of first timing. Then, at theperiod of second timing, H_(L)V_(H) and H_(H)V_(H), are transmitted onD₂ channel. As the signal is transmitted in a time division sequence,the encoder in the receiver can be arranged more simple simply.

The technique of reducing the number of the expanders in the decoderwill now be explained. FIG. 74(b) shows a time-base assignment of fourdata components 810, 810a, 810b, and 810c of a signal. When other fourdata components 811, 811a, 811b, and 811c are inserted between the fourdata components 811, 811a, 811b, and 811c respectively, the latter canbe transmitted at intervals of time. In operation, the second videodecoder 422 shown in FIG. 74(a) receives the four components of thefirst data stream D₁ at a first input 521 and transfers them through aswitch 812 to an expander 503 one after another.

More particularly, the component 810 first fed is expanded during thefeeding of the component 811 and after completion of processing thecomponent 810, the succeeding component 810a is fed. Hence, the expander503 can process a row of the components at time intervals by the sametime division manner as of the mixer, thus substituting for thesimultaneous operation of a number of expanders.

FIG. 75 is a time-base assignment of data components of an HDTV signal,in which H_(L)V_(L)(1) of an NTSC component of the first channel signalfor a TV program is allocated to a data domain 821 of D₁ signal. Also,H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) carrying HDTV additionalcomponents of the first channel signal are allocated to three domains821a, 821b, and 821c of D₂ signal respectively. There are provided otherdata components 822, 822a, 822b, and 822c between the data components ofthe first channel signal which can thus be expanded with an expandercircuit during transmission of the other data. Hence, all the datacomponents of one channel signal will be processed by a single expandercapable of operating at a higher speed.

Similar effects will be ensured by assignment of the data components toother domains 821, 821a, 821b, and 821c as shown in FIG. 76. Thisbecomes more effective in transmission and reception of a common 4 PSKor ASK signal having no different digital levels.

FIG. 77 shows a time-base assignment of data components during physicaltwo-level transmission of three different signal level data: e.g. NTSCHDTV, and super HDTV or low resolution NTSC, standard resolution NTSC,and HDTV. For example, for transmission of three data components of lowresolution NTSC, standard NTSC, and HDTV, the low resolution NTSC orH_(L)V_(L) is allocated to the data domain 821 of D₁ signal. Also,H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) of the standard NTSC componentare allocated to three domains 821a, 821b, 821c respectively.H_(L)V_(H)-H, H_(H)V_(L)-H, and H_(H)V_(H)-H, of the HDTV component areallocated to domains 823, 823a, and 823b respectively.

Here, as shown by the block diagrams of FIGS. 156 and 170, a logic levelarrangement based on discrimination in the error correction capabilityas described in the second embodiment is added to 4 VSB or 8 VSB. Moreparticularly, H_(L)V_(L) is carried on D₁₋₁ channel of the D₁ signal.The D₁₋₁ channel is higher in the error correction capability than D₁₋₂channel, as described in the second embodiment. The D₁₋₁ channel ishigher in the redundancy but lower in the error rate than the D₁₋₂channel and the date data 821 can be reconstructed at a lower C/N ratethan that of the other data 821a, 821b, and 821C. More specifically, alow resolution NTSC component will be reproduced at a far location fromthe transmitter antenna or in a signal attenuating or shadow area, e.g.the interior of a vehicle.

In view of the error rate, the data 821 of D₁₋₁ channel is less affectedby signal interference than the other data 821a, 821b, and 821c of D₁₋₂channel, while being specifically discriminated and stayed in adifferent logic level, as described in the second embodiment. While D₁and D₂ are divided into two physically different levels, the levelsdetermined by discrimination of the distance between error correctingcodes are arranged differently in the logic level.

The demodulation of D₂ data requires a higher C/N rate than that for D₁data. In action, H_(L)V_(L) or low resolution NTSC signal can at leastbe reproduced in a distant or lower C/N service area. H_(L)V_(H),H_(H)V_(L), and H_(H)V_(H) can in addition be reproduced at a lower C/Narea. Then, at a high C/N area, H_(L)V_(H)-H, H_(H)V_(L)-H, andH_(H)V_(H)-H components can also be reproduced to develop an HDTVsignal. Accordingly, three different level broadcast signals can beplayed back. This method allows the signal receivable area shown in FIG.53 to increase from a double region to a triple region, as shown in FIG.90, thus ensuring a higher opportunity for enjoying TV programs.

FIGS.FIG. 78 is a block diagram of the third video decoder arranged forthe time-base assignment of data shown in FIG. 77, which is similar tothat shown in FIG. 72 except that the third input 551 for D₃ signal iseliminated and the arrangement shown in FIG. 74(a) is added.

In operation, both the D₁ and D₂ signals are fed through two input units521 and 530 respectively to a switch 812 at the first timing. As theircomponents including H_(L)V_(L) are time divided, they are transferredin a sequence by the switch 812 to an expander 503. This sequence willnow be explained referring to the time-base assignment of FIG. 77. Acompressed form of H_(L)V_(L) of the first channel is first fed to theexpander 503 where it is expanded. Then, H_(L)V_(H), H_(H)V_(L), andH_(H)V_(H) are expanded. All the four expanded components are sentthrough a switch 812a to a mixer 556 where they are mixed to produceH_(L)V_(L)-H. H_(L)V_(L)-H is then fed back from the terminal 1 of aswitch 765a through the input 2 of a switch 765 to the H_(L)V_(L) inputof the mixer 556.

At the second timing, H_(L)V_(H)-H, H_(H)V_(L)-H, and H_(H)V_(H)-H ofthe D₂ signal shown in FIG. 77 are fed to the expander 503 where theyare expanded before transferred through the switch 821a to the mixer556. They are mixed by the mixer 556 to an HDTV signal which is fedthrough the terminal 2 of the switch 765a to the output unit 521 forfurther delivery. The time-base assignment of data components fortransmission, shown in FIG. 77, contributes to the simplest arrangementof the expander and mixer. Although FIG. 77 shows two, D₁ and D₂, signallevels, four-level transmission of a TV signal will be feasible usingthe addition of a D₃ signal and a super resolution HDTV signal.

FIG. 79 illustrates a time-base assignment of data components of aphysical three-level, D₁, D₂, D₃, TV signal, in which data components ofthe same channel are so arranged as not to overlap with one another withtime. FIG. 80 is a block diagram of a modified video decoder 423,similar to FIG. 78, in which a third input 521a is added. The time-baseassignment of data components shown in FIG. 79 also contributes to thesimple construction of the decoder.

The operation of the modified decoder 423 is almost identical to thatshown in FIG. 78 and associated with the time-base assignment shown inFIG. 77 and will not be explained in greater detail. It is also possibleto multiplex data components on the D₁ signal as shown in FIG. 81.However, two data components 821 and 822 have increased error correctioncapability than the other data components 821a, 812b, and 812c, thusstaying at a higher signal level. More particularly, the data assignmentfor transmissions made in one physical level but two logic levelrelationship. Also, each data component of the second channel isinserted between two adjacent data components of the first channel sothat serial processing can be executed at the receiver side and the sameeffects as of the time-base assignment shown in FIG. 79 will thus beobtained.

The time-base assignment of data components shown in FIG. 81 is based onthe logic level mode and can also be carried in the physical level modewhen the bit transmission rate of the two data components 821 and 822 isdecreased to ½ or ⅓ thus to lower the error rate. The physical levelarrangement consists of three different levels.

FIG. 82 is a block diagram of another modified video decoder 423 fordecoding of the D₁ signal time-base arranged as shown in FIG. 81, whichis simpler in construction than that shown in FIG. 80. Its operation isidentical to that of the decoder shown in FIG. 80 and will not beexplained in greater details.

As understood, the time-base assignment of data components shown in FIG.81 also contributes to the similar arrangement of the expander andmixer. Also, four data components of the D₁ signal are fed at respectivetime slices to a mixer 556. Hence, the circuitry arrangement of themixer 556 or a plurality of circuit blocks such as provided in the videomixer 548 of FIG. 32 may be arranged for changing the connectiontherebetween corresponding to each data component so that they becomecompatible in time division operation and thus, minimized in circuitryconstruction.

Accordingly, the receiver can be minimized in the overall construction.

It would be understood that the fifth embodiment is not limited to ASKmodulation and the other methods including PSK and QAM modulation, suchas described in the first, second, and third embodiments, will beemployed with equal success.

Also, FSK modulation will be eligible in any of the embodiments. Forexample, the signal points of a multiple-level FSK signal consisting offour frequency components F1 f1, f2, f3, and f4 are divided into groupsas shown in FIG. 58 and when the distance between any two groups arespaced from each other for ease of discrimination, the multi-leveltransmission of the FSK signal can be implemented, as illustrated inFIG. 83.

More particularly, it is assumed that the frequency group 841 of f1 andf2 is assigned D₁=0 and the group 842 of f3 and f4 is assigned D₁=1. Iff1 and f3 represent 0 at D₂ and f2 and f4 represent 1 at D₂, two-bitdata transmission, one bit at D₁ or D₂, will be possible as shown inFIG. 83. When the C/N rate is high, a combination of D₁=0 and D₂=1 isreconstructed at t=t3 and a combination of D₁=1 and D₂=0 at t=t4. Whenthe C/N rate is low, D₁=0 only is reproduced at t=t3 and D₁=1 at t=t4.In this manner, the FSK signal can be transmitted in the multi-levelarrangement. This multi-state FSK signal transmission is applicable toeach of the third, fourth, and fifth embodiments.

The fifth embodiment may also be implemented in the form of a magneticrecord/playback apparatus of which block diagram shown in FIG. 84because its ASK mode action is appropriate to magnetic record andplayback operation.

The effects of television broadcasting using the 8-level VSB shown inFIGS. 68(a), (b), and (c) are described below.

While the transmitted data quantity is high with 8-level VSB, it alsohas a higher error rate than 4-level VSB for the same C/N value.However, in high image quality HDTV transmissions, the availabletransmission capacity makes it possible to apply more error correctioncoding, and the error rate can thus be reduced. This band capacity alsoenable enables multi-level (hierarchical) television broadcasts andother new features in the future.

The relative effects of 4-level, 8-level, and 16-level VSB are describedbelow.

In ground station broadcasts using the NTSC or PAL frequency band, theusable transmission band is effectively limited to approximately 5 MHzbecause of the 6-MHz frequency limit of the NTSC format, for example, asshown in FIG. 136. With 4-level VSB, the effective data transmissionquantity is 5 MHz×4=20 Mbps because the frequency utilization efficiencyis 4 bits/Hz. A minimum of 15 Mbps-18 Mbps is required, however, fordigital HDTV signal transmission. Because there is no spare capacitywith 4-level VSB, the redundancy used for error correction is only10-20% of the HDTV effective transmission quantity as shown in thecomparison chart in FIG. 169.

With 8-level VSB, the effective data transmission quantity is 5 MHz×6=30Mbps because the frequency utilization efficiency is 5 bits/Hz. While 15Mbps-18 Mbps is required for digital HDTV signal transmission asdescribed above, when using 8-level VSB modulation, more than 50% of theactual HDTV signal transmission quantity can be used for errorcorrection coding as shown in FIG. 169. As shown by error rate curves805 and 806 in FIG. 161 163, the error rate relative to the same C/Nvalue in the transmission system is less with TCM 8-level VSB than with4-level VSB, even through error correction code gain is greater with8-level VSB than with 4-level VSB, because significantly more errorcorrection coding can be added with 8-level VSB during ground stationbroadcasting of same-data-rate HDTV digital signals using the 6-MHzband. As a result, 8-level VSB with high code gain error correctioncoding also has the effect of enabling a larger service area for groundstation HDTV broadcasts than does 4-level VSB. While the increased sizeof the error correction circuits required with 8-level VSB does increasethe complexity of the receiver circuitry, the circuit scale of theequalizer in the receiver is significantly smaller than that ofreceivers using QAM modulation, which contains a phase component,because VSB and ASK are amplitude modulation methods. As a result, an8-level VSB circuit board containing the error correction circuit issmaller than an equivalent 32-level QAM board with the same transmissioncapacity.

A digital HDTV receiver with an appropriate circuit scale and a largeground station broadcasting service area can therefore be achieved using8-level VSB.

Note that the ECC 744a and trellis encoder 744b in the transmitter andreceiver block diagrams of FIG. 84 for the present embodiment, FIGS.131, 137, 156, and 157 for embodiment 6, and FIG. 144 for embodiment 9are used as examples of the specific error correction method, and the4-, 8-, and 16-level VSB modulator 749 described with reference to FIG.61 are used for transmission. The VSB demodulator 760 described withreference to FIG. 63 is used in the receiver to generate the digitalreception data by means of the 4-, 8-, and 16-level level slicer 757from the 4-, 8-, and 16-level VSB signal. After error correction bymeans of the trellis decoder 759b and ECC decoder 759a, described belowwith reference to FIG. 84 for the present embodiment, and FIGS. 131,137, 156, and 157 for embodiment 6, a digital HDTV signal is generatedby the image expander of the image decoder 402, and the digital HDTVsignal is then output.

As shown in FIGS. 160(a) and (b) described below with the sixthembodiment, the ECC encoder 744a uses a Reed-Solomon encoder 744j andinterleaver 744k, and uses a deinterleaver 759k and Reed-Solomon decoder759j for the ECC decoder 759a. Applying interleaving as described in theprevious embodiment improves resistance to transmission system noisesuch as burst error.

Code gain can be further increased and the error rate decreased by usinga trellis encoder as shown in FIGS. 128(a), (b), (c), (d), (e), and (f).A ratio 2/3 trellis encoder 744b 743c and decoder 759b as shown in FIG.172 are most appropriate with 8-level VSB because of 3 bits/symbolcoding. The data quantity is compressed 2/3 in this case.

The embodiments have been described using primarily the example of amultilevel (hierarchical) digital television signal. While an idealbroadcasting format can be achieved using a multilevel signal, the imagecompression circuit and modulator/demodulator circuits become morecomplex, and are therefore not preferable due to cost for the start ofnew broadcasting services. As described at the beginning of the fifthembodiment, a broadcasting system with a simple television circuit canbe achieved by using a signal-signal interval L=L₀, i.e., an equalinterval, in the 4-level VSB and 8-level VSB signals and anon-multilevel television transmission, and by simplifying the circuitshown in FIG. 137 as shown in FIG. 157. Once the HDTV format is incommon use, it is then possible to change to a hierarchical 8-level VSBtransmission format.

Four- and 8-level VSB have been described above, and 16- and 32-levelVSB are described below with reference to FIGS. 159(a)-(b). FIG. 159(a)shows the 16-level VSB constellation. As shown in FIG. 159(b), thesignal between two signal points is grouped into eight groups 722a-722h,which are treated as eight signal points and can be treated as 8-levelVSB signals to enable a two-stage multilevel transmission. In this case,multilevel transmission can be achieves achieved with time divisionmultiplexing even when intermittently transmitting an 8-level VSBsignal. The maximum data rate with this method is ⅔. In FIG. 157(c)159(c), the data is further grouped into four groups 723a-723d, whichcan be handled as 4-level VSB signals adding one more level to thehierarchy. While the maximum data rate drops with time divisionmultiplex transmission of 4-level VSB signals, multilevel transmissionis possible with 3-stage multilevel VSB transmission.

With this method, a multilevel transmission whereby 8-level VSB or4-level VSB data can be reproduced when the C/N ratio of the 16-levelVSB data deteriorates can be achieved. By doubling the signal points ofthe 16-level VSB format as shown in FIG. 159(d), 32-level VSBtransmission is enabled. When 16-level VSB capacity is increased in thefuture, this method will maintain compatibility while making it possibleto obtain a 6-bit/symbol data capacity.

By summarizing the above, the VSB receiver shown in the block diagram ofFIG. 161 and the VSB transmitter shown in the block diagram of FIG. 162can be achieved.

While 4-level VSB and 8-level VSB are used by way of example above,16-level VSB as shown in FIGS. 159(a)-(c) can also be used fortransmission. With 16-level VSB, a 40-Mbps transmission capacity can beused with a 6-MHz band in ground station broadcasting. Because the datarate of the HDTV digital compression signal is 15-18 Mbps using the MPEGstandard, there is excessive reserve in the transmission capacity. Asshown in FIG. 169, redundancy R₁₆=100% or greater; redundancy istherefore too great for transmitting one channel digital HDTV, and thecircuitry is simply made more complex with little additional advantagegained over 8-level VSB. In addition, 16-level VSB redundancy is onlyabout 10%, the same as 4-level VSB redundancy, in ground station HDTVbroadcasting of two programs with 16-level VSB. As a result, the servicearea is reduced because sufficient error correction coding cannot beapplied with two-program 16-level VSB. As described before, sufficienterror correction cannot be applied with 4-level VSB because theredundancy R₄=10-20% and the service area is limited. As will be knownfrom FIG. 169, sufficient error correction coding can be achieved with8-level VSB because the redundancy R₈=50%. A broad service area can alsobe achieved without significantly.

FIG. 84 will now be explained in more detail. FIG. 174 is a blockdiagram showing a circuitry arrangement of QAM/VSB compatible modulatorfor multi-level transmission according to Embodiment 5. The input 742comprises a first data stream 743 and a second data stream 744 which arecombined by a processor 745 and then code allocated to I-channel andQ-channel by an I/Q mapping 1000 prior to quadrature modulation. For theVSB modulation, the I/Q mapping 1000 delivers data on one or I-axis ofthe two channels while the I and Q channels are the same in the level.For the QAM modulation, the code assignment is executed according to theconstellation diagram. Resultant mapped signal outputs of the I/Qmapping 1000 are transmitted to two FIR filters 1001 and 1002 where theyare weighted with roll-off characteristics before being supplied to DCoffsets 1003 and 1004 respectively. In the VSB system, the tapcoefficient values of the FIR filters 1001 and 1002 are determined sothat two filter outputs are orthogonal to each other. More particularly,a pair of equal input data streams are converted by the FIR filters 1001and 1002 to two discrete code forms which are plotted in orthogonalrelationship to each other in the coordinate. Also, in the VSB system, aportion of the carrier is allowed to pass through for ease ofreproduction at the receiver side. This is done at the DC offsets 1003and 1004. The tap coefficients of the FIR filters 1001 and 1002 areselected with a VSB/QAM selector 1009 for determining the type ofmodulation. A control output of the QAM/VSB selector 1009 actuates acoefficient generator 1008 for producing and delivering givencoefficients to the two FIR filters 1001 and 1002. Two outputs of the DCoffsets 1003 and 1004 are fed to a data selector 1005 where their I/Qcomponents are alternately selected to produce a VSB or QAM modulatedsignal. The resultant digital modulated signal of the data selector 1005is converted by a D/A converter to its analog form which is furthertransmitted from an output 748. The action of the data selector 1005 iscontrolled by a four-time symbol frequency clock produced and suppliedfrom a clock generator 1007.

As explained, the foregoing compatible hardware arrangement is capableof producing both QAM and VSB modulated signals by selectivelydetermining the coefficient values of the coefficient generator 1008.

FIG. 175 is a block diagram showing another circuitry arrangement of theQAM/VSB modulator for multi-level transmission according to Embodiment5. Two, first and second, data streams are multiplexed by a processor745 and code allocated by an I/Q mapping 1000 to the I and Qcoordinates. Resultant two, I and Q, channel data outputs of the I/Qmapping 1000 are transferred to an I/Q selector 1010 where one of the Iand Q signals is selected using a four-time symbol frequency clockproduced and supplied from a clock generator 1007. The selected channelmodulated signal is waveform shaped by an FIR filter 1011,digital-to-analog converted by a D/A converter 1012, and furthertransmitted through an output 748.

In this arrangement, the filter circuit is simple as uniaxial althoughthe filter processing speed has to be increased to two times that of theprevious arrangement.

FIG. 176 illustrates a third modification of the QAM/VSB modulator ofEmbodiment 5. Two, first and second, data streams are multiplexed by aprocessor 745 and transferred to an I/Q mapping 1000. I and Q channelsignals plotted to their respective code points by the I/Q mapping 1000are subjected to VSB orthogonal impulse response processing of two FIRfilters 1026 and 1027 and transmitted to two multipliers 1020 and 1021respectively. The multipliers 1020 and 1021 are loaded with differentwaveform data determined by a cos table 1024 and a sin table 1023respectively to which a quasi-carrier is supplied from a counter 1025.Accordingly, the I and Q channel signals are multiplied with the cos andsin signals respectively at their respective multipliers 1020 and 1021.Two modulated outputs of the multipliers 1020 and 1021 are fed to anadder 1022 where their I and Q data are combined. A resultant compositesignal is then converted by a D/A converter 1006 to its analog formwhich is further delivered from an output 748.

In the third modification, the carrier frequency of the analog modulatedsignal can be changed by varying the counting interval of the counter1025.

FIG. 177 is a block diagram showing a Trellis decoder in thedemodulator. The Trellis encoderdecoder 759b explained with FIG. 84 maychange the coding graingain according to the level of the transmitterside and the conditions of a transmission line. Hence, a correspondingTrellis decoder is needed in the receiver side.

The performance of the Trellis decoder depends on how a code path isdetermined through internal calculation and should be associated withsome path memories.

The results of calculation with the Trellis decoder denoted by 1030 arestored in a given number of path memories, generally 1032, 1033, and1034, merged in a path memory group 1031. The number of the results ispreliminarily determined from the needs at the receiver side, dependingon e.g. the transmission channel basis or the broadcasting system of atype. For the purpose, an address control signal is supplied to a memoryaddress generator 1035 which in turn addresses a corresponding one ofthe path memories to select a code path.

This arrangement allows the single Trellis decoder 1030 associated withthe plural memories to perform multiple characteristics as acting as anumber of the Trellis decoders. Thus, the demodulator will be reduced inthe hardware construction as compared with a conventional circuitarrangement where a plurality of the Trellis decoders are coupled inparallel for selective use.

FIG. 178 is a block diagram of a receiver for interception of VSBmulti-level transmitted signals emitted in the air. It is also essentialfor digital broadcasting service to eliminate any signal interferencewhich is common in the conventialconventional analog broadcastingsystems.

For elimination of such an interference, notch filters have been usedwith optimum success. However, any fixed notch filter can hardly conducta frequency offset signal used in the conventional analog broadcastingsystem. The receiver shown in FIG. 178 offers an improvement. An analogor digital signal from an antenna 32a is selectively received by a tuner752 and data extracted by an I/Q detector 1040. During the detection, afrequency error correction signal is produced by an AFC detector 1043from orthogonal data and fed back to the tuner 752. The AFC detector1043 is adapted to vary the feedback of the frequency error correctionsignal using an external control. The interference by an analog signalin a detected signal is examined by an interference detector 1041 whereit is compared with stored interference patterns for identification. Ifthe intercepted digital signal carries a cochannel interference by ananalog signal, it is subjected to a notch filter characteristic actionof an NTSC rejection filter 1042. If not, the digital signal is directlypassed without the rejecting action of the filter 1042. When theinterference detector 1041 has found an increase of the interference inthe intercepted signal resulting from the frequency offset at thetransmitter side, it causes a carrier offset detector 1044 to produceand transmit a command signal via the AFC detector 1043 to the tuner 752for changing its local oscillation frequency. As the result, theinterference level measured at the interference detector 1041 isdecreased to a point which represents the optimum receiving level of thereceiver. An output signal of the NTSC rejection filter 1042 is then fedto an equalizer 1045 where its ghost component of removed. In addition,a BER (bit error ratio) counter 1048 is provided for detecting remainingerrors in the equalized signal and if so, the signal is fed back foroptimizing the signal interception in relation to the frequency offset.Finally, the equalized signal from the equalizer 1045 is transferred toa rotator 1046 for carrier reproduction and then to a decision 1047 forreproduction of data from the corresponding code points.

FIG. 179 illustrates another arrangement of the QAM/VSB compatiblereceiver.

Intercepted broadcasting waveforms including QAM and VSB signals are fedto a tuner 1050. After tuning, an intermediate frequency signal of theselected waveform is band rejected by a filter (SAW) 1051 and convertedby an analog converter 1052 to an A/D converter enable frequency. Whenthe selected waveform is a VSB signal, it is subjected to uniaxialdetection and AFC detection of a VSB detector 1054. An A/D converter1053 digitizes a QAM modulated signal directly and a baseband signalextracted from the VSB signal.

When the digitized signal is a QAM signal, it is subjected to digitalAFC action of an AFC 1055, I/Q data reconstruction of a QAM detector1056, and waveform shaping of a roll-off filter 1057. The waveformshaped signal is then transmitted to a waveform equalizer 1059 forremoval of unwanted ghost. The output signal of the equalizer 1059 isphase compensated by a carrier recovery 1060 and code reconstructed by adecision 1061. Meanwhile, an AGC detector 1064 produces an AGC signaland an AFC detector 1062 calculates a frequency change from the outputof the carrier recovery 1060.

A code form signal of the decision 1061 is then transmitted through adeinterleaver 1065, a Trellis decoder 1066, a deinterleaver 1067, and aReed-Solomon decoder 1068 and released after an error correction.

The VSB signal output of the A/D converter 1053 is directly fed to theequalizer 1059 and then processed in the same manner as for the QAMsignal. It is also subjected to the same error correction as the QAMsignal.

There is a clock recovery 1058 operable with the entire symbols for theQAM signal. For the VSB signal, a clock reference signal of a burst formis extracted by a gate signal generator from the VSB output signal ofthe VSB detector 1054 and fed to the clock recovery 1058 for actuationof its PLL.

EMBODIMENT 6

A sixth embodiment of the present invention is a magnetic recording andplayback apparatus in which the above transmission and recording methodis employed. Although in the fifth embodiment a multiple-level ASK datatransmission is described, but it is also feasible in the same manner toadopt this invention in a magnetic recording and playback apparatus of amulti-level ASK recording system as shown in the block diagram of FIG.173. A multi-level or non-multilevel magnetic recording can be realizedby applying the C-CDM method of the present invention to PSK, FCK, andQAM, as well as ASK.

First of all, the method of realizing a multi-level recording in a 16QAM or 32 QAM magnetic recording playback apparatus will be explained incompliance with the C-CDM method of the present invention. FIG. 84 is acircuit block diagram showing a 16 QAM 32 QAM, 4 ASK, 8 ASK, 16 ASK, 8PSK system incorporating C-CDM modulator. Hereinafter, a QAM systembeing multiplexed by the C-CDM method is termed as SRQAM. FIGS. 137 and154 show block diagrams in which SRQAM is applied to the transmissionsystem, such as broadcast.

ASAs shown in FIG. 84, an input video signal, e.g. an HDTV signal, to amagnetic record/playback apparatus 851 is divided and compressed by avideo encoder 401 into a low frequency band signal through a first videoencoder 401a and a high frequency band signal through a second videoencoder 401b respectively. Then, a low frequency band component, e.g.H_(L)V_(L), of the video signal is fed to a first data stream input 743of an input unit 742 and a high frequency band component includingH_(H)V_(H) is fed to a second data stream input 744 of the same. The twocomponents are further transferred to a modulator 749 of amodulator/demodulator unit 852 852a. The first data stream input 743adds an error correcting code to the low frequency band signal in an ECC743a. On the other hand, the second data stream fed into the second datastream input 744 is 2 bit in case of 16 SRQAM, 3 bit in case of 36SRQAM, and 4 bit in case of 64 SRQAM. After an error correcting code isencoded by an ECC 744a, this signal is supplied to a Trellis encoder744b, such as shown in FIGS. 128(a), (b), (c) in which a Trellis encodedsignal having a ratio ½ in case of 16 SRQAM, 2/3 in case 32 SRQAM, and3/4 in case of 64 SRQAM, is produced. A 64 SRQAM signal, for example,has a first data stream of 2 bit and a second data stream of 4 bit. ATrellis encoder 744b of FIG. 128(c) allows this 64 SRQAM signal toperform a Trellis encoding of ratio 3/4 wherein 3 bit data is convertedinto 4 bit data. In the case of 4 ASK, 8 ASK, and 16 ASK, the Trellisencoding at the ratio of 1/2, 2/3, and 4/3 can be carried ourout solely.Thus, redundancy increases and a data rate decreases, while errorcorrecting capability increases. This results in the reduction of anerror rate-inrate in the same data rate. Accordingly, transmittableinformation amount of the recording/playback system or transmissionsystem will increase substantially.

Since the 8 VSB transmission system described above in connection withthe fifth embodiment takes 3 bits per symbol, the Trellis encoder 744gand the Trellis decoder 774q with the ratio 2/3 as shown in FIGS.128(b), (e) can be used, and the entire block diagram will be as shownin FIG. 171.

It is, however, possible to constitute the first data stream input 743not to include a Trellis encoder as shown in FIG. 84 of this sixthembodiment because the first data stream has low error rate inherently.This will be advantageous in view of the simplification of circuitconfiguration. The second data stream, however, has a narrow inter-codedistance as compared with the first data stream and, therefore, has aworse error rate. The Trellis encoding of the second data streamimproves such a worse error rate. It is no doubt that an overall circuitconfiguration becomes simple if the Trellis encoding of the first datastream is eliminated. An operation for modulation is almost identical tothat of the transmitter of the fifth embodiment shown in FIG. 64 andwill be not be explained in greater detail. A modulated signal of themodulator 749 is fed into a recording/playback circuit 853 in which itis AC biased by a bias generator 856 and amplified by an amplifier 857a.Thereafter, the signal is fed to a magnetic head 854 854b for recordingonto a magnetic tape 855.

A format of the recorded signal is shown in a recording signal frequencyassignment of FIG. 113. A main, e.g. 16 SRQAM, signal 859 having acarrier of frequency fc records information, and also a pilot f_(p)signal 859a having a frequency 2fc is recorded simultaneously.Distortion in the recording operation is lowered as a bias signal 859bhaving a frequency 2fc is recorded simultaneously. Distortion in therecording operation is lowered as a bias signal 859b having a frequencyf_(BIAS) adds AC bias for magnetic recording. Two of three-level signalsshown in FIG. 113 are recorded in multiple state. In order to reproducethese recorded signals, two thresholds Th-1-2, Th-2 are given. A signal859 will reproduce all of two levels while a signal 859 will reproduceD₁ data only, depending on C/N level of the recording/playback.

A main signal of 16 SRQAM will have a signal point assignment shown inFIG. 10. Furthermore, a main signal of 36 SRQAM will have a signal pointassignment shown in FIG. 100. When 4 ASK, 8 ASK are used, theconstellation will be as shown in FIGS. 58, 68(a) and (b). Inreproduction of this signal, both the main signal 859 and the pilotsignal 859a are reproduced through the magnetic head 854 854a andamplified by an amplifier 857b. An output signal of the amplifier 857bis fed to a carrier reproduction circuit 858 in which a filter 858aseparates the frequency of the pilot signal f_(p) having a frequency 2f02f₀ and ½ frequency divider 858b reproduces a carrier of frequency f0 f₀to transfer it to a demodulator 760. This reproduced carrier is used todemodulate the main signal in the demodulator 760. Assuming that amagnetic recording tape 855, e.g. HDTV tape, is of a high C/N rate, 16signal points are discriminatable and thus both D₁ and D₂ aredemodulated in the demodulator 760. Subsequently, a video decoder 402reproduce reproduces all the signals. An HDTV VCR can reproduce a highbit-rate TV signal such as 15 Mbps HDTV signal. The low lower the C/Nrate is, the cheaper the cost of a video tape is. So far, a VHS tape inthe market is inferior more than 10 dB in C/N rate to a full-scalebroadcast tape. If a video tape 855 is of low C/N rate, it will not beable to discriminate all the 16 or 32 valued signal points. Thereforethe first data stream D₁ can be reproduced, while a 2 bit, 3 bit, or 4bit data stream of the second data stream D₂ cannot be reproduced. Only2 bit data stream of the first data stream is reproduced. If a two-levelHDTV video signal is recorded and reproduced, a low C/N tape havinginsufficient capability of reproducing a high frequency band videosignal can output only a low rate low frequency band video signal of thefirst data stream, specifically e.g. a 7 Mbps wide NTSC TV signal.

As shown in a block diagram of FIG. 144, a second data stream output759, the second data stream input 744, and the second video decoder 402acan be eliminated in order to provide customers one aspect of lowergrade products. In this case, a recording/playback apparatus 851,dedicated to a low bit rate, will include a modulator such as amodulated QPSK which modulates or demodulates the first date data streamonly. This apparatus allows only the first data stream to be recordedand reproduced. Specifically, a wide NTSC grade video signal can berecorded and reproduced.

Above-described high C/N rate video tape 855 capable of recording a highbit-rate signal, e.g. HDTV signal, will be able to use be used in such alow bit-rate dedicated magnetic recording/playback apparatus but willreproduce the first data stream D₁ only. That is, the wide NTSC signalis outputted, while the second data stream is not reproduced. In otherwords, one recording/playback apparatus having a complicatedconfiguration can reproduce a an HDTV signal and the otherrecording/playback apparatus having a simple configuration can reproducea wide NTSC signal if a given video tape 855 includes the samemulti-level HDTV signal. Accordingly in case of two-level multiplestate, four combinations will be realized with prefect perfectcompatibility among two tapes having different C/N rates and tworecording/playback apparatus having different recording/playback datarates. This will bring a remarkable effect. In this case, an NTSCdedicated apparatus will be simple in construction as compared with anHDTV dedicated apparatus. In more detail, a circuitry scale of ECTV EDTVdecoder will be ⅙ of that of HDTV decoder. Therefore, a low functionapparatus can be realized at fairly low cost. Realization of two, HDTVand EDTV, types recording/playback apparatus having differentrecording/reproducing capability of picture quality will provide varioustype products ranging in a wide price range. Users can freely select atape among a plurality of tapes from an expensive high C/N rate tape toa cheaper low C/N rate tape, as occasion demands so as to satisfyrequired picture quality. Not only maintaining perfect compatibility butobtaining expandable capability will be attained and furthercompatibility with a future system will be ensured. Consequently, itwill be possible to establish long-lasting standards forrecording/playback apparatus. Other recording methods will be used inthe same manner. For example, a multi-level recording will be realizedby use of phase modulation explained in the first and third embodiments.A recording using ASK explained in the fifth embodiment will also bepossible. A two or three multi-level state will be realized byconverting present recording from two-level to four-level ASK or toeight-level ASK and dividing into two group as shown in FIGS. 59(c) and59(d) or in FIGS. 68(a) and 68(b).

A circuit block diagram for ASK will be as is shown in FIG. 173 which issimilar to that disclosed in FIG. 84. By the combination of Trellis andASK, the error rate will be reduced. Besides embodiments alreadydescribed, a multi-level recording will be also realized by use ofmultiple tracks on a magnetic type tape. Furthermore, a theoreticalmulti-level recording will be feasible by differentiating the errorcorrecting capability so as to discriminate respective data.

Compatibility with future standards will be described below. A settingof standards for recording/playback apparatus such as VCR is normallydone by taking account of the most highest C/N rate tape available inpractice. The recording characteristics of a tape progresses rapidly.For example, the C/N rate has been improved more than 10 dB comparedwith the tape used 10 years ago. If it is supposed that new standardswill be established after 10 to 20 years due to an advancement of tapeproperty, a conventional method will encounter with difficulty inmaintaining compatibility with older standards. New and old standards,in fact, used to be one-way compatible or non-compatible with eachother. On the contrary, in accordance with the present invention, thestandards are first of all established for recording and/or reproducingthe first data stream and/or second data stream on present day tapes.Subsequently, if the C/N rate is improved magnificently in future, anupper level data stream, e.g. a third data stream, will be added withoutany difficulty as long as the present inventions incorporated in thesystem. For example, a super HDTV VCR capable of recording orreproducing three-level 64 SRQAM or 8 ASK will be realized whilemaintaining perfect compatibility with the conventional standards. Amagnetic tape, recording first to third data streams in compliance withnew standards, will be able to use, of course, in the older two-levelmagnetic recording/playback apparatus capable of recording and/orreproducing only first and second data streams. In this case, first andsecond data streams can be reproduced perfectly although the third datastream is left non-reproduced. Therefore, an HDTV signal can bereproduced. For these reasons, the merit of expanding recording dataamount while maintaining compatibility between new and old standards isexpected.

Returning to the explanation of reproducing operation of FIG. 84, themagnetic head 854 854a and the magnetic reproduction circuit 853 858reproduce a reproducing signal from the magnetic tape 855 and feeds feedit to the modulation/demodulation circuit 852 a demodulator unit 852b.The demodulating operation is almost identical with that of first,third, and fourth embodiments and will no further be explained. Thedemodulator 760 reproduces the first and second data stream D₁ and D₂.The second data stream D₂ is error corrected with high code gain in aTrellis-decoder 759b such as Vitabi decoder, so as to be low error rate.The video decoder 402 demodulates D₁ and D₂ signals to output an HDTVvideo signal.

FIG. 131 is block diagram showing a three-level magneticrecording/playback apparatus in accordance with the present inventionwhich includes one theoretical level in addition to two physical levels.This system is substantially the same as that of FIG. 84. The differenceis that the first data stream is further divided into two subchannels byuse of a TDM in order to realize a three-level constitution.

As shown in FIG. 131, an HDTV signal is separated first of all into two,medium and low frequency band video signals D₁₋₁ and D₁₋₂, through a 1-1video encoder 401c and 1-2 video encoder 401d and, thereafter, fed intoa first data stream input 743 of an input section 742. The data streamD₁₋₁ having a picture quality of MPEG grade is error correcting codedwith high code gain in an ECC coder 743a, while the data stream D₁₋₂ iserror correcting coded with normal code gain in an ECC encoder 743b.D₁₋₁ and D₁₋₂ are time multiplexed together in a TDM 743c to be one datastream D1. D₁ and D₂ are modulated into two-level signals in a C-CDM 749and then recorded on the magnetic tape 855 through the magnetic head854.

In playback operation, a recording signal reproduced through themagnetic head 854 is demodulated into D₁ and D₂ by the C-CDM demodulator760 in the same manner as in the explanation of FIG. 84. The first datastream D₁ is demodulated into two, D₁₋₁ and D₁₋₂, subchannels throughthe TDM 758c provided in the first data stream output 758. D₁₋₁ data iserror corrected in an ECC decoder 758a having high code gain. Therefore,D₁₋₁ data can be demodulated at a lower C/N rate as compared with D₁₋₂data. A 1-1 video decoder 402a decodes the D₁₋₁ data and outputs an LDTVsignal. On the other hand, D₁₋₂ data is error corrected in an ECCdecoder 75 8b 758b having normal code gain. Therefore, D₁₋₂ data has athreshold value of high C/N rate compared with D₁₋₁ data and thus willnot be demodulated when a signal level is not large D₁₋₂ data is thendemodulated in a 1-2 video decoder 4 02d 402d and summed with D₁₋₁ datato output an EDTV signal of wide NTSC grade.

The second data stream D₂ is Vitabi demodulated in a Trellis decoder759b and error corrected at an ECC decoder 7 59a 759a. Thereafter, D₂data is converted into a high frequency band video signal through asecond video decoder 402b and, then, summed with D₁₋₁ and D₁₋₂ data tooutput an HDTV signal. In this case, a threshold value of the C/N rateof D₂ data is set larger than that of C/N rate of D₁₋₂ data.Accordingly, D₁₋₁ data, i.e. an LDTV signal, will be reproduced from atape 855 having a smaller C/N rate. D₁₋₁ and D₁₋₂ data, i.e. an EDTVsignal, will be reproduced from a tape 855 having a normal C/N rate.And, D₁₋₁, D₁₋₂, and D₂, i.e. an HDTV signal, will be reproduced from atape 855 having a high C/N rate.

Three-level magnetic recording/playback apparatus can be realized inthis manner. As described in the foregoing description, the tape 855 hasan interrelation between C/N rate and cost. The present invention allowsusers to select a grade of tape in accordance with a content of TVprogram they want to record because video signals having picturequalities of three grades are recorded and/or reproduced in accordancewith tape cost.

Next, an effect of multi-level recording will be described with respectto fast feed playback. As shown in a recording track diagram of FIG.132, a recording track 855a having an azimuth angle A and a recordingtrack 855b having an opposite azimuth angle B are alternately arrayed onthe magnetic tape 855. The recording track 855a has a recording region855c at its central portion and the remainder as D₁₋₂ recording regions855d, as denoted in the drawing. This unique recording pattern isprovided on at least one of several recording tracks. The recordingregion 855c records one frame of LDTV signal. A high frequency bandsignal D₂ is recorded on a D₂ recording region 855e corresponding to anentire recording region of the recording track 855a. This recordingformat causes no novel effect against a normal speed recording/playbackoperation.

A fast feed reproduction in a reverse direction does not allow amagnetic head trace 855f having an azimuth angle A to coincide with themagnetic track as shown in the drawing. As the present inventionprovides the D₁₋₁ recording region 8 55c 855c at a central narrow regionof the magnetic tape as shown in FIG. 132, this region only is surelyreproduced although it occurs at a predetermined probability. Thusreproduced D₁₋₁ signal can demodulate an entire picture plane of thesame time although its picture quality is an LDTV of MPEG1 level. Inthis manner several to several tens LDTV signals per second can bereproduced with perfect picture images during the fast feed playbackoperation, thereby enabling users to surely confirm picture imagesduring the fast feed operation.

A head trace 855g corresponds to a head trace in the reverse playbackoperation, from which it is understood only a part of the magnetic trackis traced in the reverse playback operation. The recording/playbackformat shown in FIG. 132 however allows, even in such a reverse playbackoperation, to reproduce D₁₋₁ recording region and, therefore, ananimation of LDTV grade is outputted intermittently.

Accordingly, the present invention makes it possible to record a pictureimage of LDTV grade within a narrow region on the recording track, whichresults in intermittent reproduction of almost perfect still pictureswith picture quality of LDTV grade during normal and reverse fast feedplayback operations. Thus, the users can easily confirm picture imageseven in high-speed seaching.

Next, another method will be described to respond a higher speed fastfeed playback operation. A D₁₋₁ recording region 855c is provided asshown at lower right of FIG. 132, so that one frame of LDTV signal isrecorded thereon. Furthermore, a narrow D₁₋₁-D₂ recording region 855h isprovided at a part of the D₁₋₁ recording region 855c. A subchannel D₁₋₁in this region records a part of information relating to the one frameof LDTV signal. The remainder of the LDTV information is recorded on theD₂ recording region 855j of the D₁₋₁.D₂ recording region 855h in aduplicated manner. The subchannel D₂ has a data recording capacity 3 to5 times as much as the subchannel D₁₋₁. Therefore, subchannels D₁₋₁ andD₂ can record one frame information of LDTV signal on a smaller, 1/3_(?)/5 1/3˜ 1/5, area of the recording tape. As the head trace can berecorded in a further narrower regions 855h, 855j, both time and areaare decreased into 1/3 _(?)/5 1/3˜ 1/5 as compared with a head tracetime T_(S1). Even if the trace of head is further inclined by increasingfast feed speed amount, the probability of entirely tracing this regionwill be increased. Accordingly, perfect LDTV picture images will beintermittently reproduced even if the fast feed speed is increased up to3 to 5 times as fast as the case of the subchannel D₁₋₁ only.

In case of a two-level VCR, this method is useless in reproducing the D₂recording region 855j and therefore this region will not be reproducedin a high-speed fast feed playback operation. On the other hand, athree-level high performance VCR will allow users to confirm a pictureimage even if a fast feed playback operation is executed at a faster, 3to 5 times as fast as two-level VCR, speed. In other words, not onlyexcellent picture quality is obtained in accordance with the cost but amaximum fast feed speed capable of reproducing picture images can beincreased in accordance with the cost.

Although this embodiment utilizes a multi-level modulation system, it isneedless to say that a normal, e.g. 16 QAM, modulation system can alsobe adopted to realize the fast feed playback operation in accordancewith the present invention as long as an encoding of picture images isof multiple type.

A recording method of a conventional non-multiple digital VCR, in whichpicture images are highly compressed, disperses video data uniformly.Therefore, it was not possible in a fast feed playback operation toreproduce all the picture images on a picture plane of the same time.The picture reproduced was the one consisting of a plurality of pictureimage blocks having non-coincident time bases. The present invention,however, provides a multi-level HDTV VCR which can reproduce pictureimage blocks having coincided time bases on a picture plane during afast feed playback operation although its picture quality is of LDTVgrade.

The three-level recording in accordance with the present invention willbe able to reproduce a high resolution TV signal such as HDTV signalwhen the recording/playback system has a high C/N rate. Meanwhile, a TVsignal of EDTV grade, e.g. a wide NTSC signal, or a TV signal of LDTVgrade, e.g. a low resolution NTSC signal, will be outputted when therecording/playback system has a low C/N rate or poor function.

As it is described in the foregoing description, the magneticrecording/playback apparatus in accordance with the present inventioncan reproduce picture images consisting of the same content even if C/Nrate is low or error rate is high, although the resolution or thepicture quality is relatively low.

EMBODIMENT 7

A seventh embodiment of the present invention will be described forexecution of four-level video signal transmission. A combination of thefour-level signal transmission and the four-level video dataconstruction will create a four-level signal service area as shown inFIG. 91. The four-level service area is consisted of, from innermost, afirst 890a, a second 890b, a third 890c, and a fourth signal receivingarea 890d. The method of developing such a four-level service area willbe explained in more detail.

The four-level arrangement can be implemented by using four physicallydifferent levels determined through modulation or four logic levelsdefined by data discrimination in the error correction capability. Theformer provides a large difference in the C/N rate between two adjacentlevels and the C/N rate has to be increased to discriminate all the fourlevels from each other. The latter is based on the action ofdemodulation and a difference in the C/N rate between two adjacentlevels should stay at minimum. Hence, the four-level arrangement is bestconstructed using a combination of two physical levels and two logiclevels. The division of a video signal into four signal levels will beexplained.

FIG. 93 is a block diagram of a divider circuit 3 which comprises avideo divider 895 and four compressors 405a, 405b, 405c, and 405d. Thevideo divider 895 contains three dividers 404a, 404b, and 404c which arearranged identical to the divider circuit 404 of the first video encoder401 shown in FIG. 30 and will be not be explained in greater detail. Aninput video signal is divided by the dividers into four components,H_(L)V_(L) of low resolution data, H_(H)V_(H) of high resolution data,and H_(L)V_(H) and H_(H)V_(L) for medium resolution data. The resolutionof H_(L)V_(L) is a half that of the original input signal.

The input video signal is first divided by the divider 4 04a 404a intotwo, high and low, frequency band components, each component beingdivided into two, horizontal and vertical, segments. The intermediatebetween the high and low frequency ranges is a dividing point accordingto the embodiment. Hence, if the input video signal is an HDTV signal of1000 line vertical resolution, H_(L)V_(L) has a vertical resolution of500 lines and a horizontal resolution of a half value.

Each of two, horizontal and vertical, data of the low frequencycomponent H_(L)V_(L) is further divided by the divider 404c into twofrequency band segments. Hence, an H_(L)V_(L) segment output is 250lines in the vertical resolution and ¼ of the original horizontalresolution. This output of the divider 404c which is termed as an LLsignal is then compressed by the compressor 405a to a D₁₋₁ signal.

The other three higher frequency segments of H_(L)V_(L) are mixed by amixer 772c to an LH signal which is then compressed by the compressor405b to a D₁₋₂ signal. The compressor 405b may be replaced by threecompressors provided between the divider 404c and the mixer 772c.

H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) form from the divider 404a aremixed by a mixer 772a to an H_(H)V_(H)-H signal. If the input signal isas high as 1000 lines in both horizontal and vertical resolution,H_(H)V_(H)-H has 500 to 1000 lines of a horizontal and a verticalresolution. H_(H)V_(H)-H is fed to the divider 404b where it is dividedagain into four components.

Similarly, H_(L)V_(L) from the divider 404b has 500 to 750 lines of ahorizontal and a vertical resolution and transferred as an H_(L) signalto the compressor 405c. The other three components, H_(L)V_(H),H_(H)V_(L), and H_(H)V_(H), from the divider 404b have 750 to 1000 linesof a horizontal and a vertical resolution and are mixed by a mixer 772bto an HH signal which is then compressed by the compressor 405d anddelivered as a D₂₀₂ D₂₋₂ signal. After compression, the HL signal isdelivered as a D₂₋₁ signal. As the result, LL or D₁₋₁ carries afrequency data of 0 to 250 lines, LH or D₁₋₂ carries a frequency datafrom more than 250 lines up to 500 lines, HL or D₂₋₁ carries a frequencydata of more than 500 lines up to 750 lines, and HH or D₂₋₂ carries afrequency data of more than 750 lines to 1000 lines so that the dividercircuit 3 can provide a four-level signal. Accordingly, when the dividercircuit 3 of the transmitter 1 shown in FIG. 87 is replaced by thedivider circuit of FIG. 93, the transmission of a four-level signal willbe implemented.

The combination of multi-level data and multi-level transmission allowsa video signal to be at steps declined in the picture quality inproportion to the C/N rate during transmission, thus contributing to theenlargement of the TV broadcast service area. At the receiving side, theaction of demodulation and reconstruction is identical to that of thesecond receiver of the second embodiment shown in FIG. 88 and will notbe explained in greater detail. In particular, the mixer 37 is modifiedfor video signal transmission rather than data communications and willnow be explained in more detail.

As described in the second embodiment, a received signal after beingdemodulated and error corrected, is fed as a set of four componentsD₁₋₁, D₁₋₂, D₂₋₁, and D₂₋₂ to the mixer 37 of the second receiver 33 ofFIG. 88.

FIG. 94 is a block diagram of a modified mixer 33 in which D₁₋₁, D₁₋₂,D₂₋₁, and D₂₋₂ are explained by their respective expanders 523a, 523b,523c, and 523d to an LL, andan LH, an HL, and an HH signal respectivelywhich are equivalent to those described with FIG. 93. If the bandwidthof the input signal is 1, LL has a bandwidth of ¼, LL+LH has a bandwidthof ½, LL+LH+HL has a bandwidth of ¾, and LL+LH+HL+HH has a bandwidthof 1. The LH signal is then divided by a divider 531a and mixed by avideo mixer 548a with the LL signal. An output of the video mixer 548ais transferred to an H_(L)V_(L) terminal of a video mixer 548c. Thevideo mixer 531a is identical to that of the second decoder 527 of FIG.32 and will not be explained in greater detail. Also, the HH signal isdivided by a divider 531b and fed to a video mixer 548b. At the videomixer 548b, the HH signal is mixed with the HL signal to an H_(H)V_(H)-Hsignal which is then divided by a divider 531c and sent to the videomixer 548c. At the video mixer 548c, H_(H)V_(H)-H is combined with thesum signal of LH and LL to a video output. The video output of the mixer33 is then transferred to the output unit 36 of the second receivershown in FIG. 88 where it is converted to a TV signal for delivery. Ifthe original signal has 1050 lines of vertical resolution or is an HDTVsignal of about 1000-line resolution, its four different signal levelcomponents can be intercepted in their respective signal receiving areasshown in FIG. 91.

The picture quality of the four different components will be describedin more detail. The illustration of FIG. 92 represents a combination ofFIGS. 86 and 91. As apparent, when the C/N rate increases, the overallsignal level of amount of data is increased from 862d to 862a by stepsof four signal levels D₁₋₁, D₁₋₂, D₂₋₁, D₂₋₂.

Also, as shown in FIG. 95, the four different level components LL, LH,HL, and HH are accumulated in proportion to the C/N rate. Morespecifically, the quality of a reproduced picture will be increased asthe distance from a transmitter antenna becomes small. When L=Ld, LLcomponent is reproduced. When L=Lc, LL+LH signal is reproduced. WhenL=Lb, LL+LH+HL signal is reproduced. When L=La, LL+LH+HL+HH signal isreproduced. As the result, if the bandwidth of the original signal is 1,the picture quality is enhanced at ¼ increments of bandwidth from ¼ to 1depending on the receiving area. If the original signal is an HDTV of1000-line vertical resolution, a reproduced TV signal is 250, 50 0 500,750, and 1000 lines in the resolution at their respective receivingareas. The picture quality will thus be varied at steps depending on thelevel of a signal. FIG. 96 shows the signal propagation of aconventional digital HDTV signal transmission system, in which no signalreproduction will be possible when the C/N rate is less than V0. Also,signal interception will hardly be guaranteed at signal interferenceregions, shadow regions, and other signal attenuating regions, denotedby the symbol x, of the service area. FIG. 97 shows the signalpropagation of an HDTV signal transmission system of the presentinvention. As shown, the picture quality will be a full 1000-line gradeat the distance La where C/N=a, a 750-line grade at the distance Lbwhere C/N=b, a 500-line grade at the distance Lc where C/N=c, and a250-line grade are the distance Ld where C/N=d. Within the distance La,there are shown unfavorable regions where the C/N rate drops sharply andno HDTV quality picture will be reproduced. As understood, a lowerpicture quality signal can however be intercepted and reproducedaccording to the multi-level signal transmission system of the presentinvention. For example, the picture quality will be a 750-line grade atthe point B in a building shadow area, a 250-line grade at the point Din a running train, a 750-line grade at the point F in a ghostdeveloping area, a 250-line grade at the point G in a running car, a250-line grade at the point L in a neighbor signal interference area. Asset forth above, the signal transmission system of the present inventionallows a TV signal to be successfully received at a grade in the areawhere the conventional system is poorly qualified, thus increasing itsservice area. FIG. 98 shown shows an example of simultaneousbroadcasting of four different TV programs, in which three qualityprograms C, B, A are transmitted on their respective channels D₁₋₂,D₂₋₁, D₂₋₂ while a program D identical to that of a local analogue TVstation is propagated on the D₁₋₁ channel. Accordingly, while theprogram D is kept available at simulcast service, the other threeprograms can also be distributed on air for offering a multiple programbroadcast service.

EMBODIMENT 8

Hereinafter, an eighth embodiment of the present invention will beexplained referring to the drawings. The eighth embodiment employs amulti-level signal transmission system of the present invention for atransmitter/receiver in a cellular telephone system.

FIG. 115 is a block diagram showing a transmitter/receiver of a portabletelephone, in which a telephone conversation sound inputted across amicrophone 76 2 762is compressed and coded in a compressor 405 intomulti-level, D₁, D₂, and D₃, data previously described. These D₁, D₂,and D₃ data are time divided in a time division circuit 765 intopredetermined time slots and, then, modulated in a modulator 4 into amulti-level, e.g. SRQAM, signal previously described. Thereafter, anantenna sharing unit 764 and an antenna 22 transmit a carrier wavecarrying a modulated signal, which will be intercepted by a base stationlater described and further transmitted to other base stations or acentral telephone exchanger so as to communicate with other telephones.

On the contrary, the antenna 22 receives transmission radio waves fromother base stations as communication signals from other telephones. Areceived signal is demodulated in a multiple-level, e.g. SRQAM, typedemodulator 45 into D₁, D₂, and D₃ data. A timing circuit 767 detectstiming signals on the basis of demodulated signals. These timing signalsare fed into the time division circuit 765. Demodulated signals D₁, D₂,and D₃ are fed into an expander 503 and expanded into a sound signal,which are transmitted to a speaker 763 and converted into sound.

FIG. 116 shows a block diagram exemplarily showing an arrangement ofbase stations, in which three base stations 7 71 771, 772, and 773locate at center of respective receiving cells 768, 769, and 770 ofhexagon or circle. These base stations 771, 772, and 773 respectivelyhashave a plurality of transmitter/receiver units 761a˜761j each similarto that of FIG. 115 so as to have data communication channels equivalentto the number of these transmitter/receiver units. A base stationcontroller 774 is connected to all the base stations and always monitorsa communication traffic amount of each base station. Based on themonitoring result, the base station controller 774 carries out anoverall system control including allocation of channel frequencies torespective base stations or control of receiving cells of respectivebase stations.

FIG. 117 is a view showing a traffic distribution of communicationamount in a conventional, e.g. QPSK, system. A diagram d=A shows data774a and 774b having frequency utilization efficiency 2 bit/Hz, and adiagram d=B shows data 774c of frequency utilization efficiency 2bit/Hz. A summation of these data 774a, 774b, and 774c becomes a data774d, which represents a transmission amount of Ach consisting ofreceiving cells 768 and 770. Frequency utilization efficiency of 2bit/Hz is uniformly distributed. However, density of population in anactual urban area is locally high in several crowded areas 775a, 775b,and 775c which includes buildings concentrated. A dataData 774erepresenting a communication traffic amount shows several peaks atlocations just corresponding to these crowded areas 775a, 775b, and775c, in contrast with other areas having a small communication amount.A capacity of conventional cellular telephone was uniformly set to 2bit/Hz frequency efficiency at entire region as shown by the data 774dirrespective of actual traffic amount TF shown by the data 774e. It isnot not effective to give the same frequency efficiency regardless ofactual traffics amount. In order to compensate for this ineffectiveness,the conventional systems have allocated many frequencies to the regionshaving a large traffic amount, increased channel number, or decreasedthe receiving cells thereof. However, an increase of channel number isrestricted by the frequency spectrum. Furthermore, conventionalmulti-level; e.g. 16 QAM or 64 QAM, mode transmission systems increasetransmission power. A reduction in the receiving cells will induce anincrease in number of base stations, thus increasing installation cost.

It is ideal for the improvement of an overall system efficiency toincrease the frequency efficiency of the region having a larger trafficamount and decrease the frequency efficiency of the region having asmaller traffic amount. A multi-level signal transmission system inaccordance with the present invention realizes this ideal modification.This will be explained with reference to FIG. 118 showing acommunication amount & and traffic distribution in accordance with theeighth embodiment of the present invention.

More specifically, FIG. 118 shows communication amounts of respectivereceiving cells 770b, 768, 769, 770, and 770a taken along a line A-A′.The receiving cells 768 and 770 utilize frequencies of a channel groupA, while the receiving cells 770b, 769, and 770a utilize frequencies ofa channel group B which does not overlap with the cannel channel groupA. The base station controller 774 shown in FIG. 116 increases ordecreases the channel number of these channels in accordance with thetraffic amount of respective receiving cells. In FIG. 118, a diagram d=Arepresents a distribution of a communication amount of the A channel. Adiagram d=B represents a distribution of a communication amount of the Bchannel. A diagram d=A+B represents a distribution of a communicationamount of all the channels. A diagram TF represents a communicationtraffic amount, and a diagram P shows a distribution of buildings andpopulation.

The receiving cells 768, 769, and 770 employ the multi-level, e.g.SRQAM, signal transmission system. Therefore, it is possible to obtain afrequency utilization efficiency of 6 bit/Hz, three times as large as 2bit/Hz of QPSK, in the vicinity of the base stations as denoted by data776a, 776b, and 776c. Meanwhile, the frequency utilization efficiencydecreases at steps from 6 bit/Hz to 4 bit/Hz, and 4 bit/Hz to 2 bit/Hz,as it goes to suburban area. if If the transmission power isinsufficient, 2 bit/Hz areas become narrower than the receiving cells,denoted by dotted lines 777a, 777b, and 777c, of QPSK. However, anequivalent receiving cell will be easily obtained by slightly increasingthe transmission power of the base stations.

Transmitting/receiving operation of a mobile station capable ofresponding to a 64 SRQAM signal is carried out by use of modified QPSK,which is obtained by set setting a shift amount of SRQAM to S=1, at theplace far from the base station, by use of 16 SRQAM at a place not sofar from the same, and 64 SRQAM at the near place. Accordingly, themaximum transmission power does not increase as compared with QPSK.

Furthermore, 4 SRQAM type transmitter/receiver, whose circuitconfiguration is simplified as shown in a block diagram of FIG. 121,will be able to communicate with other telephones while maintainingcompatibility. That will be the same in 16 SRQAM typetransmitter/receiver shown in a block diagram of FIG. 122. As a result,three different type telephones having different modulation systems willbe provided. Small in size and light in weight is important for portabletelephones. In this regard, the 4 SRQAM system having a simple circuitconfiguration will be suitable for the users who want a small and lighttelephone although its frequency utilization efficiency is low andtherefore the cost of a call may increase. In this manner, the presentinvention system can suit a wide variety of usage.

As is explained above, the transmission system having a distributionlike d=A+B of FIG. 118, whose capacity is locally altered, isaccomplished. Therefore, an overall frequency utilization efficiencywill be much effectively improved if layout of base stations isdetermined to fit for the actual traffic amount denoted by TF.Especially, effect of the present invention will be large in a microcell system, whose receiving cells are smaller and therefore numeroussub base stations are required. Because a large number of sub basestations can be easily installed at the place having a large trafficamount.

Next, data assignment of each time slot will be explained referring toFIG. 119, wherein FIG. 199(a) shows a conventional time slot and FIG.119(b) shows a time slot according to the eighth embodiment. Theconventional system performs a down, i.e. from a base station to aterminal station, transmission as shown in FIG. 119(a), in which a syncsignal S is transmitted by a time slot 780a and transmission signals torespective terminal stations of A, B, C channels by time slots 780b,780c, and 780d respectively at a frequency A. On the other hand, an up,i.e. from the mobile station to the base station, transmission isperformed in such a manner that a sync signal S, and transmissionsignals of a, b, and c channels are transmitted by time slots 781a,781b, 781c, and 781d at a frequency B.

The present invention, which is characterized by a multi-level, e.g. 64SRQAM, signal transmission system, allows to have three-level dataconsisting of D₁, D₂, and D₃ of 2 bit/Hz as shown in FIG. 119(b). Asboth the A₁ and A₂ data are transmitted by 16 SRQAM, their time slotshave two times the data rate as shown by slots 782b and 782c and 783band 783c. It means the same quality sound can be transmitted half thetime. Accordingly, a time width of respective time slots 782 782b and782c becomes halved. In this manner, two times the transmission capacitycan be acquired at the two-level region 776c shown in FIG. 118, i.e. atthe vicinity of the base station.

In the same way, time slots 782g and 783g carry out thetransmission/reception of E1 data by use of a 64 SRQAM signal. As thetransmission capacity is three times, one time slot can be used forthree channels of E₁, E₂, and E₃. This would be used for an area furtherclose to the base station. Thus, up to three times the communicationcapacity can be obtained at the same frequency band. An actualtransmission efficiency, however, would be reduced to 90%. It isdesirable for enhancing the effect of the present invention to coincidethe transmission amount distribution according to the present inventionwith the regional distribution of the actual traffic amount as perfectas possible.

In fact, an actual urban area consists of a crowded building districtand a greenbelt zone surrounding this building area. Even an actualsuburb area consists of a residential district and fields or a forestsurrounding this residential district. These urban and suburb areasresemble the distribution of the TF diagram. Thus, the application ofthe present invention will be effective.

FIG. 120 is a diagram showing time slots by the TDMA method, whereinFIG. 120(a) shows a conventional method and FIG. 120(b) shows thepresent invention. The conventional method uses time slots 786a and 786bfor transmission to portable phones of A and B channels at the samefrequency and time slots 787a and 787b for transmission from the same,as shown in FIG. 120(a).

On the contrary, 16 SRQAM mode of the present invention uses a time slot788a for reception of A₁ channel and a time slot 788c for transmissionto A1 channels as shown in FIG. 1 20(b) 120(b). A width of the time slotbecomes approximately ½. In case of 64 SRQAM mode, a time slot 788i isused for reception of D₁ channel and a time slot 7881 is used fortransmission to D₁ channel. A width of the time slot becomesapproximately ⅓.

In order to save electric power, a transmission of E₁ channel isexecuted by use of a normal 4 SRQAM time slot 788 r while reception ofE1 channel is executed by use of a 16 SRQAM time slot 788p being a ½time slot. Transmission power is surely suppressed, althoughcommunication cost may increase due to a long occupation time. This willbe effective for a small and light portable telephone equipped with asmall battery or when the battery is almost worn out. As is described inthe foregoing description, the present invention makes it possible todetermine the distribution of transmission capacity so as to coincidewith an actual traffic distribution, thereby increasing substantialtransmission capacity. Furthermore, the present invention allows basestations or terminal stations to freely select one among two or threetransmission capacities. If the frequency utilization efficiency islowered, power consumption will be decreased. If the frequencyutilization efficiency is selected higher, communication cost will besaved. Moreover, adoption of a 4 SRQAM having smaller capacity willsimplify the circuitry and reduce the size and cost of the telephone. Asexplained in the previous embodiments, one characteristics of thepresent invention is that compatibility is maintained among all ofassociated stations. In this manner, the present invention not onlyincreases transmission capacity but allows to provide customers a widevariety of series from a super mini telephone to a high performancetelephone.

EMBODIMENT 9

Hereinafter, a ninth embodiment of the present invention will bedescribed referring to the drawings. The ninth embodiment employs thisinvention in an OFDM transmission system. FIG. 123 is a block diagram ofa an OFDM transmitter/receiver, and FIG. 124 is a diagram showing aprinciple of an OFDM action. An OFDM is one of FDM and has a betterefficiency in frequency utilization as compared with a general FDM,because an OFDM sets adjacent two carriers to be quadrate with eachother. Furthermore, OFDM can bear multipath obstructions such as ghostsand, therefore, may be applied in the future to the digital musicbroadcasting or digital TV broadcasting.

As shown in the principle diagram of FIG. 124, OFDM converts an inputsignal by a serial to parallel converter 791 into a data being disposedon a frequency axis 793 at intervals of 1/ts, so as to producesubchannels 794a⁻ 94e 794a-794e. This signal is inversely FFT convertedby a modulator 4 having an inverse FFT 40 into a signal on a time axis79 ⁻ 9 799 to produce a transmission signal 795. This inverse FFT signalis transmitted during an effective symbol period 796 of the time periodts. A guard interval 797 having an amount tg is provided between symbolperiods.

A transmitting/receiving action of HDTV signal in accordance with thisninth embodiment will be explained referring to the block diagram ofFIG. 123, which shows a hybrid OFDM-CCDM system. An inputted HDTV signalis separated by a video encoder 401 into three-level three-levels, a lowfrequency band D₁₋₁, a medium-low frequency band D₁₋₂, and ahigh-medium-low frequency band D₂, video signals, and fed into an inputsection.

In a first data stream input 743, D₁₋₁ signal is ECC encoded with highcode gain and D₁₋₂ signal is ECC coded with a normal code gain. A TDM743 performs time division multiplexing of D₁₋₁ and D₁₋₂ signals toproduce a D₁ signal, which is then fed to a D₁ serial to parallelconverter 791d in a modulator 852a. D₁ signal consists of n pieces ofparallel data, which are inputted into first inputs of n pieces of C-CDMmodulator 4a, 4b,—respectively.

On the other hand, the high frequency band signal D₂ is fed into asecond data stream input 744 of the input section 742, in which D₂signal is ECC (Error Correction Code) encoded in an ECC 744a and thenTrellis encoded in a Trellis encoder 744b. Thereafter, the D₂ signal issupplied to a D₂ serial to parallel converter 791b of the modulator 852aand converted into n pieces of parallel data, which are inputted intosecond inputs of the n pieces of C-CDM modulator 4a, 4b,—respectively.

The C-CDM modulators 4a, 4b, 4c—respectively produces produce 16 SRQAMsignal on the basis of D₁ data of the first data stream input and D₂data of the second data stream input. These n pieces of C-CDM modulatorrespectively has have a carrier different from each other. As shown inFIG. 124, carriers 79 4a 794a, 794b, 794c,—are arrayed on the frequencyaxis 793 so that adjacent two carriers are 90°-out-of-phase with eachother. Thus C-CDM modulated n pieces of modulated signal are fed intothe inverse FFT circuit 40 and mapped from the frequency axis dimension793 to the time axis dimension 790 799. Thus, time signals 796a, 796b—,having an effective symbol length ts, are produced. There is provided aguard interval zone 797a of Tg seconds between the effective symbol timezones 796a and 796b, in order to reduce multipath obstruction. FIG. 129is a graph showing a relationship between time axis and signal level.The guard time Tg of the guard interval band 797a is determined bytaking account of multipath affection and usage of signal. By settingthe guard time Tg longer than the multipath affected time, e.g. TVghost, modulated signals from the inverse FFT circuit 40 are convertedby a parallel to serial converter 4e into one signal and, then,transmitted from a transmitting circuit 5 as an RF signal.

Next, an action of a receiver 43 will be described. A received signal,shown as time-base symbol signal 796e of FIG. 124, is fed into an inputsection 24 of FIG. 123. Then, the received signal is converted into adigital signal in a demodulator 852b and further changed into Fouriercoefficients in a FFT 40a. Thus, the signal is mapped from the time axis799 to the frequency axis 793a as shown in FIG. 124. That is, thetime-base symbol signal is converted into frequency-base carriers 794a,794b,—. As these carriers are in quadrature relationship with eachother, it is possible to separate respective modulated signals. FIG.125(b) shows thus demodulated 16 SRQAM signal, which is then fed torespective C-CDM demodulators 45a, 45b,—of a C-CDM demodulator 45, inwhich demodulated 16 SRQAM signal is demodulated into multi-level subsignals D₁, D₂. These sub signals D₁ and D₂ are further demodulated by aD₁ parallel to serial converter 852a and a D₂ parallel to serialconverter 852b into original D₁ and D₂ signals.

Since the signal transmission system is of C-CDM multi-level shown in125(b), both D₁ and D₂ signals will be demodulated under betterreceiving condition but only D₁ signal will be demodulated under worse,e.g. low C/N rate, receiving condition. Demodulated D₁ signal isdemodulated in an output section 757. As D₁₋₁ signal has higher ECC codegain as compared with the D₁₋₂ signal, an error signal of the D₁₋₁signal is reproduced even under worse receiving condition.

The D₁₋₁ signal is converted by a 1-1 video decoder 402c into a lowfrequency band signal and outputted as an LDTV, and the D₁₋₂ signal isconverted by a 1-2 video decoder 402d into a medium frequency bandsignal and outputted as EDTV.

The D₂ signal is Trellis decoded by a Trellis decoder 75 9b 759b andconverted by a second video decoder 402b into a high frequency bandsignal and outputted as an HDTV signal. Namely, an LDTV signal isoutputted in case of the low frequency band signal only. An EDTV signalof a wide NTSC grade is outputted if the medium frequency band signal isadded to the low frequency band signal, and an HDTV signal is producedby adding low, medium, and high frequency band signals. As well as theprevious embodiment, a TV signal having a picture quality depending on areceiving C/N rate can be received. Thus, the ninth embodiment realizesa novel multi-level signal transmission system by combining an OFDM anda C-CDM, which was not obtained by the OFDM alone.

An OFDM is certainly strong against multipath such as TV ghost becausethe guard time Tg can absorb an interference signal of multipath.Accordingly, the OFDM is applicable to the digital TV broadcasting forautomotive vehicle TV receivers. Meanwhile, no OFDM signal is receivedwhen the C/N rate is less than a predetermined value because its signaltransmission pattern is non not of a multi-level type.

However the present invention can solve this disadvantage by combiningthe OFDM with the C-CDM, thus realizing a graditional gradationaldegradation depending on the C/N rate in a video signal receptionwithout being disturbed by multipath.

When a TV signal is received in a compartment of a vehicle, not only thereception is disturbed by multipath but the C/N rate is deteriorated.Therefore, the broadcast service area of a TV broadcast station will notbe expanded as expected if the countermeasure is only for multipath.

On the other hand, a reception of TV signal of at least LDTV grade willbe ensured by the combination with the multi-level transmission C-CDMeven if the C/N rate is fairly deteriorated. As a picture plane size ofan automotive vehicle TV is normally less than 10 inches, a TV signal ofan LDTV grade will provide a satisfactory picture quality. Thus, theLDTV grade service area of automotive vehicle TV will be largelyexpanded. If an OFDM is used in an entire frequency band of HDTV signal,present semiconductor technologies cannot prevent circuitry scale fromincreasing so far.

Now, an OFDM method of transmitting only D₁₋₁ of low frequency band TVsignal will be explained below. As shown in a block diagram in FIG. 138,a medium frequency band component D₁₋₂ and a high frequency bandcomponent D₂ of an HDTV signal are multiplexed in C-CDM modulator 4a,and then transmitted at a frequency band A through an FDM 40d.

On the other hand, a signal received by a receiver 43 is first of allfrequency separated by an FDM 40e and, then, demodulated by a C-CDMdemodulator 4b of the present invention. Thereafter, thus C-CDMdemodulated signal is reproduced into medium and high frequencycomponents of HDTV in the same way as in FIG. 123. An operation of avideo decoder 402 is identical to that of embodiments 1, 2, and 3 andwill not be explained in greater detail.

Meanwhile, the D₁₋₁ signal, a low frequency band signal of MPEG 1 gradeof HDTV, is converted by a serial to parallel converter 791 into aparallel signal and fed to an OFDM modulator 852c, which executes QPSKor 16 QAM modulation. Subsequently, the D₁₋₁ signal is converted by aninverse FFT 40 into a time-base signal and transmitted at a frequencyband B through a FDM 40d.

On the other hand, a signal received by the receiver 43 is frequencyseparated in the FDM 40e and then converted into a number offrequency-base signals in an FFT 40a of an OFDM modulator demodulator852d. Thereafter, frequency-base signals are demodulated in respectivedemodulators 4a, 4b,—and are fed into a parallel to serial converter882a, wherein a D₁₋₁ signal is demodulated. Thus, a D₁₋₁ signal of LDTVgrade is outputted from the receiver 43.

In this manner, only an LDTV signal is OFDM modulated in the multi-levelsignal transmission. The method of FIG. 138 makes it possible to providea complicated OFDM circuit only for an LDTV signal. A bit rate of LDTVsignal is 1/20 of that of an HDTV. Therefore, the circuit scale of theOFDM will be reduced to 1/20, which results in an outstanding reductionof overall circuit scale.

An OFDM signal transmission system is strong against multipath and willsoon be applied to a moving station, such as a portable TV, anautomotive vehicle TV, or a digital music broadcast receiver, which isexposed under strong and variable multipath obstruction. For such usagesa small picture size of less than 10 inches, 4 to 8 inches, is themainstream. It will be thus guessed that the OFDM modulation of a highresolution TV signal such as HDTV or EDTV will bring less effect. Inother words, the reception of a TV signal of LDTV grade would besufficient for an automotive vehicle TV.

On the contrary, multipath is constant at a fixed station such as a homeTV. Therefore, a countermeasure against multipath is relatively easy.Less effect will be brought to such a fixed station by OFDM unless it isin a ghost area. Using OFDM for medium and high frequency bandcomponents of HDTV is not advantageous in view of present circuit scaleof OFDM which is still large.

Accordingly, the method of the present invention, in which OFDM is usedonly for a low frequency band TV signal as sown shown in FIG. 138, canwidely reduce the circuit scale of the OFDM to less than 1/10 withoutlosing inherent OFDM effect capable of largely reducing multipleobstruction of LDTV when received at a mobile station such as anautomotive vehicle.

Although the OFDM modulation of FIG. 138 is performed only for D₁₋₁signal, it is also possible to modulate both D₁₋₁ and D₁₋₁ D₁₋₂ by OFDM.In such a case, a C-CDM two-level signal transmission is used fortransmission of D₁₋₁ and D₁₋₂. Thus, a multi-level broadcasting beingstrong against multipath will be realized for a vehicle such as anautomotive vehicle. Even in a vehicle, the gradational graduation willbe realized in such a manner that LDTV and SDTV signals are receivedwith picture qualities depending on receiving signal level or antennasensitivity.

The multi-level signal transmission according to the present inventionis feasible in this manner and produces various effects as previouslydescribed. Furthermore, if the multi-level signal transmission of thepresent invention is incorporated with an OFDM, it will become possibleto provide a system strong against multipath and to alter datatransmission grade in accordance with receivable signal level change.

FIG. 126(a) shows another method of realizing the multi-level signaltransmission system, wherein the subchannels 794a-794c of the OFDM areassigned to a first layer 801a and the subchannels 794d-794f areassigned to a second layer 801b. There is provided a frequency guardzone 802a of f_(g) between these two, first and second, layers. FIG.126(b) shows an electric power difference 802b of Pg which is providedto differentiate the transmission power of the first and second layers801a and 801b.

Utilization of this differentiation makes it possible to increaseelectric power of the first layer 801a in the range not obstructing theanalogue TV broadcast service as shown in FIG. 108(d) previouslydescribed. In this case, a threshold value of the C/N ratio capable ofreceiving the first layer 801a becomes lower than that for the secondlayer 801b as shown in FIG. 108(e). Accordingly, the first layer 801acan be received even in a low signal-level area or in a large-noisearea. Thus, a two-layer signal transmission is realized as shown in FIG.147. This is referred to as Power-Weighted-OFDM system (i.e. PW-OFDM) inthis specification. If this PW-OFDM system is combined with the C-CDMsystem previously explained, three layers will be realized as shown inFIG. 108(e) and, accordingly, the signal receivable area will becorrespondingly expanded.

FIG. 144 shows a specific circuit, wherein the first layer data passingthrough the first data stream circuit 791 a is modulated into thecarriers f₁-f₃ by the modulators 4a-4c having large amplitude and, then,are OFDM modulated in the inverse FFT 40. On the contrary, the secondlayer data passing through the second data stream circuit 791b ismodulated into the carriers f₆-f₈ by the modulators 4d-4f havingordinary amplitude and, then, are OFDM modulated in the inverse FFT 40.Then, these OFDM modulated signals are transmitted from the transmitcircuit 5.

A signal received by the receiver 43 is separated into several signalshaving carriers of f₁-f_(n)through the FFT 40a. The carriers f₁-f₃ aredemodulated by the demodulators 45a-45c to reproduce the first datastream D₁, i.e. the first layer 801a. On the other hand, the carriersf₆-f₈ are demodulated by the demodulators 45d-45f to reproduce thesecond data stream D₂, i.e. the second layer 801b.

The first layer 801a has so large electrical power that it can bereceived even in a weak-signal area. In this manner, the PW-OFDM systemrealizes the two-layer multi-level signal transmission. If this PW-OFDMis combined with the C-CDM, it will become possible to provide 3-4layers. As the circuit of FIG. 144 is identical with the circuit of FIG.123 in the remaining operations and, therefore, will not be explained ingreater detail.

Next, a method of realizing a multi-level signal transmission inTime-Weighted-OFDM (i.e. TW-OFDM) in accordance with the presentinvention will be explained. Although the OFDM System is accompaniedwith the guard time zone t_(g) as previously described, adverse affectsof ghosts will be eliminated if the delay time t_(M) of the ghost, i.e.multipath, signal satisfies the requirement of t_(M)<t_(g). The delaytime t_(M) will be relatively small, for example in the range of severalmicrosounds microseconds, in a fixed station such as a TV receiver usedfor home use. Furthermore, as its value is constant, cancellation ofghosts will be relatively easily done. On the contrary, reflected waveswill increase in case of a mobile station such as a vehicle TV receiver.Therefore, the delay time t_(M) becomes relatively large, for example inthe range of several tens microsound microsecond. Furthermore, themagnitude of t_(M) varies in response to the running movement of thevehicle. Thus, cancellation of ghosts tends to be difficult. Hence, themulti-level signal transmission is key or essential for such a mobilestation TV receiver in order to eliminate adverse affection ofmultipath.

The multi-level signal transmission in accordance with the presentinvention will be explained below. A symbol contained in the subcannelsubchannel layer A can be intensified against the ghosts by setting aguard time t_(ga) of the layer A to be larger than a guard time t_(gb)of the layer B as shown in FIG. 146. In this manner, the multi-layersignal transmission can be realized against multipath by use ofweighting of guard time. This system is referred to asGuard-Time-Weighted-OFDM (i.e. QTW-OFDM).

If the symbol number of the symbol time Ts is not different in the layerA and in the layer B, a symbol time t_(sa) of the layer A is set to belarger smaller than a symbol time t_(sb) of the layer B. With thisdifferentiation, a carrier width Δfa of the carrier A becomes largerthan a carrier width Δfb of the carrier B. (Δfa>Δfb) (Δfa<Δfb)Therefore, the error rate becomes lower in the demodulation of thesymbol of the layer A compared with the demodulation of the symbol ofthe layer B. Thus, the differentiation of the layer A and B in theweighting of the symbol time Ts can realize a two-layer signaltransmission against multipath. This system is referred to asCarrier-Spacing-Weighted-OFDM (i.e. CSW-OFDM).

By realizing the two-layer signal transmission based on the GTW-OFDM,wherein a low-resolution TV signal is transmitted by the layer A and ahigh-frequency component is transmitted by the layer B, the vehicle TVreceiver can stably receive the low-resolution TV signal regardless oftough ghost. Furthermore, the multi-level signal transmission withrespect to the C/N ratio can be realized by differentiating the symboltime t_(S) based on the CSW-OFDM between the layers A and B. If thisCSW-OFDM is combined with the GTW-OFDM, the signal reception in thevehicle TV receiver can be further stabilized. High resolution is notnormally required to the vehicle TV or the portable TV.

As the time ratio of the symbol time including a low-resolution TVsignal is small, an overall transmission efficiency will not decrease somuch even if the guard time is enlarged. Accordingly, using the GTW-OFDMof the present invention for suppressing multipath by laying emphasis onthe low-resolution TV signal will realize the multi-layer type TVbroadcast service wherein the mobile station such as the portable orvehicle TV receiver can be compatible with the stationary station suchas the home TV without substantially lowering the transmissionefficiency. If combined with the CSW-OFDM or the C-CDM as describedpreviously, the multi-layer to the C/N ratio can be also realized. Thus,the signal reception in the mobile station will be further stabilized.

An affection of the multipath will be explained in more detail. In caseof multipath 810a, 810b, 810c, and 810d having shorter delay time asshown in FIG. 145(a), the signals of both the first and second layerscan be received and therefore the HDTV signal can be demodulated. On thecontrary, in case of multipath 811a, 811b, 811c, and 811d having longerdelay time as shown in FIG. 145(b), the B signal of the second layercannot be received since its guard time t_(gb) is not sufficiently long.However, the A signal of the first layer can be received without beingbothered by the multipath since its guard time t_(ga) is sufficientlylong. As described above, the B signal includes the high-frequencycomponent of TV signal. The A signal includes the low-frequencycomponent of TV signal. Accordingly, the vehicle TV can reproduce theLDTV signal. Furthermore, as the symbol time Tsa is set larger thansymbol time Tsb, the first layer is strong against deterioration of C/Nratio.

Such a discrimination of the guard time and the symbol time is effectiveto realize two-dimensional multi-layer signal transmission of the OFDMin a simple manner. If the discrimination of guard time is combined withthe C-CDM in the circuit shown in FIG. 123, the multi-layer signaltransmission effective against both multipath and deterioration of C/Nratio will be realized.

Next, a specific example will be described below.

The smaller the D/U ratio of the receiving signal becomes, the largerthe multipath delay time T_(M) becomes. Because, the reflected waveincreases compared with the direct wave. For example, as shown in FIG.148, if the D/U ratio is smaller than 30 dB, the delay time T_(M)exceeds 30 us μs because of increase of the reflected wave. Therefore,as can be understood from FIG. 148, it will become possible to receivethe signal even in the worst condition if the Tg is set to be largerthan 50 us μs.

Accordingly, as shown in detail in FIGS. 149(a) and 149(b), three groupsof first 801a, second 801b, and third 801 c 801c layers are assigned ina 2 ms period of 1 sec TV signal. The guard times 797a, 797b, and 797c,i.e. Tga, Tgb, and Tgc, of these three groups are weighted to be, forexample, 50 microsounds microseconds, 5 microsounds microseconds, and 1microsound microsecond, respectively, as shown in FIG. 149(c). Thus,three-layer signal transmission effective to the multipath will berealized as shown in FIG. 150, wherein three layers 801a, 801b, and 801care provided.

If the GTW-OFDM is applied to ass the picture quality, it is doubtless

At the same time, the multi-layer signal transmission effective to C/Nratio can be realized. By combining the CSW-OFDM and the CSW-OFDM, atwo-dimensional multi-layer signal transmission is realized with respectto the multipath and the C/N ratio as shown in FIG. 151. As describedpreviously, it is possible to combine the CSW-OFDM and the C-CDM of thepresent invention for preventing the overall transmission efficiencyfrom being lowered. In the first, 1-2, and 1-3 layers 801a, 851a, and851az, the LDTV grade signal can be stably received by, for example, thevehicle TV receiver subjected to the large multipath T_(M) and low C/Nratio. In the second and 2-3 layers 801b and 851b, thestandard-resolution SDTV grade signal can be received by the fixed orstationary station located, for example, in the fringe of the servicearea which is generally subjected to the lower C/N ratio and ghost. Inthe third layer 801c which occupies more than half of the service area,the HDTV grade signal can be received since the C/N ratio is high andthe ghost is less because of large direct wave. In this manner, atwo-dimensional multi-layer broadcast service effective to both the C/Nratio and the multipath can be realized by the combination of theGTW-OFDM and the C-CDM or the combination of the GTW-OFDM and theCSW-C-CDM in accordance with the present invention. thus Thus, thepresent invention realizes a two-dimensional, matrix type, multi-layersignal transmission system effective to both the C/N ratio and the 194multipath, which has not ever been realized by the prior arttechnologies.

A timing chart of a three level (HDTV, SDTV, LDTV) television signal ina two-dimensional multilevel broadcast of three C/N levels and threemultipath levels is shown in FIG. 152. As shown in the figure, the LDTVsignal is positioned in slot 796a1 of the first level of level layer A,the level with the greatest resistance to multipath interference; theSDTV synchronization signal, address signal, and other important highpriority signals are positioned in slot 796a2, which has the nextgreatest resistance to multipath interference, and slot 796b1, which hasstrong resistance to C/N deterioration. The SDTV common signal, i.e.,low priority signals, and HDTV high priority signals are positioned inlevels 2 and 3 of level B. SDTV, EDTV, HDTV, and other high frequencycomponent television signals are positioned in levels 1, 2, and 3 oflevel C.

As the resistance to C/N deterioration and multipath interferenceincreases, the transmission rate drops, causing the TV signal resolutionto drop, and achieving the three-dimensional graceful degradation effectshown in FIG. 153 and unobtainable with conventional methods. As shownin FIG. 153, the three-dimensional multilevel broadcast structure of theinvention is achieved with three parameters: C/N ratio, multipath delaytime, and the transmission rate.

The present embodiment has been described using the example of atwo-dimensional multilevel broadcast structure obtained by combiningGTW-OFDM of the invention with C-CDM of the invention as previouslydescribed, or combining GTW-OFDM, CSW-C-CDM, but other two-dimensionalmultilevel broadcast structures can be obtained by combining GTW-OFDMand power-weighted OFDM, or GTW-OFDM with other C/N ratio multileveltransmission methods.

FIG. 154 is obtained by transmitting the power of carriers 794a, 794c,and 794e with less weighting compared with carriers 794b, 794d, and794f, achieving a two level power-weighted OFDM. Two levels are obtainedby power weighting carriers 795a and 795c, which are perpendicular tocarrier 794a, to carriers 795b and 795d. While a total of four levelsare obtained, the embodiment having only two levels is shown in FIG.154. As shown in the figure, because the carrier frequencies aredistributed, interference with other analog transmissions on the samefrequency band is dispersed, and there is minimal adverse effect.

By using a time positioning varying the time width of guard times 797a,797b, and 797c for each symbol 796a, 796b, and 796c as shown in FIG.155, three-level multipath multilevel transmission can be achieved.Using the time positioning shown in FIG. 155, the A-, B-, and C-leveldata is distributed on the time axis. As a result, even if burst noiseproduced at a specific time occurs, data destruction can be preventedand the TV signal can be stably demodulated by interleaving the datafrom the different layers. In particular, by interleaving with the Alevel data distributed, interference from burst noise generated by theignition systems of other vehicles can be significantly reduced inmobile TV receivers.

Block diagrams of a specific ECC encoder 744j and a specific ECC decoder749j 759j are shown in FIG. 160a and FIG. 160b, respectively. FIG. 167is a block diagram of the deinterleaver 936b. The interleave table 954processed in the deinterleave RAM 936a of the deinterleaver 936b isshown in FIG. 168a, and interleave distance L1 is shown in FIG. 168b.

Burst noise interference can be reduced by interleaving the data in thisway. By using a 4-level VSB, 8-level VSB, or 16-level VSB transmissionapparatus as described in embodiments 4, 5, and 6, respectively, andshown in the VSB receiver block diagram (FIG. 161) and the VSBtransmitter block diagram (FIG. 162), or by using a QAM or PSKtransmission apparatus as described in embodiments 1 and 2,respectively, burst noise interference can be reduced, and televisionreception with very low noise levels can be achieved in ground stationbroadcasting.

By using 3-level broadcasting by means of the method shown in FIG. 155,LDTV grade television reception by mobile receivers, including mobile TVreceivers in motor vehicles and hand-held portable television sets, canbe stabilized because level A has the effect of reducing burst noiseinterference in addition to multipath interference and C/N ratiodeterioration.

The multi-level signal transmission method of the present invention isintended to increase the utilization of frequencies but may be suitedfor not all the transmission systems since causing some type receiversto be declined in the energy utilization. It is a good idea for use witha satellite communications system for selected subscribers to employmost advanced transmitters and receivers designed for best utilizationof applicable frequencies and energy. Such a specific purpose signaltransmission system will not be bound by the present invention.

The present invention will be advantageous for use with a satellite orterrestrial broadcast service which is essential to run in the samestandards for as long as 50 years. During the service period, thebroadcast standards must not be altered but improvements will beprovided time to time corresponding to up-to-date technologicalachievements. Particularly, the energy for signal transmission willsurely be increased on any satellite. Each TV station should provide acompatible service for guaranteeing TV program signal reception to anytype receivers ranging from today's common ones to future advanced ones.The signal transmission system of the present invention can provide acompatible broadcast service of both the existing NTSC and HDTV systemsand also, ensure a future extension to match mass date datatransmission.

The present invention concerns much on the frequency utilization thanthe energy utilization. The signal receiving sensitivity of eachreceiver is arranged different differently depending on a signal statelevel to be received so that the transmitting power of a transmitterneeds not be increased largely. Hence, existing satellites which offer asmall energy for reception and transmission of a signal can best be usedwith the system of the present invention. The system is also arrangedfor performing the same standards corresponding to an increase in thetransmission energy in the future and offering the compatibility betweenold and new type receivers. In addition, the present invention will bemore advantageous for use with the satellite broadcast standards.

The multi-level signal transmission method of the present invention ismore preferably employed for terrestrial TV broadcast service in whichthe energy utilization is not crucial, as compared with satellitebroadcast service. The results are such that the signal attenuatingregions in a service area which are attributed to a conventional digitalHDTV broadcast systems are considerably reduced in extension and also,the compatibility of an HDTV receiver or display with the existing NTSCsystem is obtained. Furthermore, the service area is substantiallyincreased so that program suppliers and sponsors can appreciate moreviewers. Although the embodiments of the present invention refer to 16and 32 QAM procedures, other modulation techniques including 64, 128,and 256 QAM will be employed with equal success. Also, multiple PSK,ASK, and FSK techniques will be applicable as described with theembodiments.

A combination of the TDM with the SRQAM of the present invention hasbeen described in the above. However, the SRQAM of the present inventioncan be combined also with any of the FDM, CDMA and frequency dispersalcommunications systems.

1. A signal transmission and reception apparatus for transmitting andreceiving an n-level VSB signal, the apparatus comprising a transmitterand a receiver; said transmitter comprising: a compression means forcompressing an input video signal to a digital video compression signal;an error correction encoding means for adding an error correction codeto the digital video compression signal to produce an error correctioncoded signal; a modulation means for modulating the error correctioncoded signal to an n-level VSB modulation signal, said modulation meanscomprising a means for allocating code points along a uniaxialmodulation coordinate system, and a filter means having a plurality ofcoefficients which are a series of impulse responses defined by plottingtimebase responses to the VSB modulation signal along the in-phase axisand its orthogonal axis for filtering a series of said code pointsallocated along the uniaxial modulation coordinate system; and atransmission means for transmitting the modulation signal, and saidreceiver comprising: a means for receiving a transmitted n-level VSBmodulation signal; a demodulation means for demodulating the receivedn-level VSB modulation signal into a digital reception signal; an errorcorrection means for error correcting the digital reception signal toobtain an error-corrected digital signal; and an expanding means forexpanding the error-corrected digital signal to obtain a video outputsignal.
 2. A transmission and reception apparatus according to claim 1,wherein the error correction means comprises a trellis decoder.
 3. Atransmission and reception apparatus according to claim 2, wherein thetrellis decoder is associated with a plurality of memories which eachholds a number of selectable correct codes.
 4. A transmission andreception apparatus according to claim 1, wherein the digital receptionsignal is divided into a high priority signal and a low priority signal,and wherein said error correction means comprises a high code gain firsterror correction means and a low code gain second error correctionmeans, said first error correction means correcting the high prioritysignal.
 5. A transmission and reception apparatus according to claim 4,wherein the high priority signal carries the address data for all data.6. A transmission and reception apparatus according to claim 4, whereinthe first error correction means comprises a trellis decoder.
 7. Asignal transmission and reception apparatus according to claim 1,further comprising a band path filtering means for filtering the n-levelVSB modulation signal before being transmitted.
 8. A signal transmissionand reception apparatus for transmitting an n-level VSB signal,comprising: a compression means for compressing an input video signalinto a digital video compression signal; an error correction encodingmeans for adding an error correction code to the digital videocompression signal to produce an error correction coded signal; amodulation means for modulating the error correction coded signal to ann-level VSB modulation signal, said modulation means comprising a meansfor allocating code points along a uniaxial modulation coordinatesystem, and a filter means having a plurality of coefficients which area series of impulse responses defined by plotting timebase responses tothe VSB modulation signal along the in-phase axis and its orthogonalaxis for filtering a series of said code points allocated along theuniaxial modulation coordinate system; and a transmission means fortransmitting the modulation signal.
 9. A signal transmission accordingto claim 8, further comprising a band path filtering means for filteringthe n-level VSB modulation signal before being transmitted.
 10. A signalreceiving apparatus comprising: a tuner for receiving a transmissionsignal containing a digital modulation signal and an analog modulationsignal and for selecting the digital modulation signal using a localoscillation signal; an interference detecting means for detectinginterference caused by the analog modulation signal from the digitalmodulation signal selected by the tuner; a notch filter means responsiveto the interference detected by the interference detecting means forremoving a carrier of the analog modulation signal in a same frequencyband as a frequency band of the digital modulation signal; an errorratio calculating means for calculating a bit error ratio of an outputof the notch filter means; and an automatic frequency correcting meansfor changing a frequency of the local oscillation signal of the tuneraccording to a level of the interference detected by the interferencedetecting means and the bit error ratio calculated by the error ratiocalculating means to compensate for a frequency offset of the carrier ofthe analog modulated signal.
 11. A signal receiving apparatus accordingto claim 10, wherein the digital modulation signal is an n-level VSBmodulation signal.
 12. A signal receiving apparatus comprising: a tunerfor receiving a transmission signal containing at least one of a VSBmodulated signal and a QAM modulated signal and for selecting one of theVSB modulated signal and the QAM modulated signal to obtain a selectedsignal; an analog-to-digital converter for converting the selectedsignal into a series of digital codes; a transversal filter provided onan orthogonal axis for suppressing a transmission distortion of theseries of digital codes with respect to both orthogonal axes to obtain aseries of filtered digital codes allocated on the orthogonal axes; acarrier recovery means for phase-compensating a carrier of the filtereddigital codes allocated on the orthogonal axis outputted from thetransversal filter; and a control means for producing a control signalto extract detected codes at equal time intervals from the VSB modulatedsignal; a clock reproducing means for phase synchronizing entire codesof the QAM modulated signal when the selected signal is the QAMmodulated signal and for phase synchronizing codes of the VSB modulatedsignal intermittently at predetermined intervals when the selectedsignal is the VSB modulated signal; and a decoding means for decoding anoutput of the carrier recovery means.
 13. A signal receiving apparatuscapable of receiving a VSB modulated signal processed by digitalmodulation and a QAM modulated signal processed by digital modulation,which are a terrestrial broadcasting signal and a cable televisionsignal, respectively, said signal receiving apparatus comprising: a QAMdemodulator operable to demodulate the QAM modulated signal to a QAMdemodulated signal; a VSB demodulator operable to demodulate the VSBmodulated signal to a VSB demodulated signal; and a video decoderoperable to decode the QAM demodulated signal, and operable to decodethe VSB demodulated signal.
 14. The signal receiving apparatus inaccordance with claim 13, wherein the video decoder is an MPEG decoder.15. A signal receiving method comprising: receiving a signal with areceiver capable of receiving a VSB modulated signal processed bydigital modulation and a QAM modulated signal processed by digitalmodulation, which are a terrestrial broadcasting signal and a cabletelevision signal, respectively; demodulating the QAM modulated signalto a QAM demodulated signal if the received signal is the QAM modulatedsignal, and decoding the QAM demodulated signal with a video decoder;and demodulating the VSB modulated signal to a VSB demodulated signal ifthe received signal is the VSB modulated signal, and decoding the VSBdemodulated signal with the video decoder, wherein the video decoder iscapable of decoding the VSB demodulated signal and the QAM demodulatedsignal.
 16. The signal receiving method in accordance with claim 15,wherein the decoding is MPEG decoding.
 17. A signal receiving apparatuscapable of receiving a VSB modulated signal processed by digitalmodulation and a PSK modulated signal processed by digital modulation,which are a terrestrial broadcasting signal and a satellite broadcastingsignal, respectively, said signal receiving apparatus comprising: a PSKdemodulator operable to demodulate the PSK modulated signal to a PSKdemodulated signal; a VSB demodulator operable to demodulate the VSBmodulated signal to a VSB demodulated signal; and a video decoderoperable to decode the PSK demodulated signal, and operable to decodethe VSB demodulated signal.
 18. The signal receiving apparatus inaccordance with claim 17, wherein the video decoder is an MPEG decoder.19. A signal receiving method comprising: receiving a signal with areceiver capable of receiving a VSB modulated signal processed bydigital modulation and a PSK modulated signal processed by digitalmodulation, which are a terrestrial broadcasting signal and a satellitebroadcasting signal, respectively; demodulating the PSK modulated signalto a PSK demodulated signal if the received signal is the PSK modulatedsignal, and decoding the PSK demodulated signal with a video decodercapable of decoding VSB demodulated signals and PSK demodulated signals;and demodulating the VSB modulated signal to a VSB demodulated signal ifthe received signal is the VSB modulated signal, and decoding the VSBdemodulated signal with the decoder capable of decoding the VSBdemodulated signals and PSK demodulated signals.
 20. The signalreceiving method in accordance with claim 19, wherein the decoding isMPEG decoding.
 21. A signal receiving apparatus capable of receiving aPSK modulated signal processed by digital modulation and a QAM modulatedsignal processed by digital modulation, which are a satellitebroadcasting signal and a cable television signal, respectively, saidsignal receiving apparatus comprising: a QAM demodulator operable todemodulate the QAM modulated signal to a QAM demodulated signal; a PSKdemodulator operable to demodulate the PSK modulated signal to a PSKdemodulated signal; and a video decoder operable to decode the QAMdemodulated signal, and operable to decode the PSK demodulated signal.22. The signal receiving apparatus in accordance with claim 21, whereinthe video decoder is an MPEG decoder.
 23. A signal receiving methodcomprising: receiving a signal with a receiver capable of receiving aPSK modulated signal processed by digital modulation and a QAM modulatedsignal processed by digital modulation, which are a satellitebroadcasting signal and a cable television signal, respectively;demodulating the QAM modulated signal to a QAM demodulated signal if thereceived signal is the QAM modulated signal, and decoding the QAMdemodulated signal with a video decoder capable of PSK demodulatedsignals and QAM demodulated signals; and demodulating the PSK modulatedsignal to a PSK demodulated signal if the received signal is the PSKmodulated signal, and decoding the PSK demodulated signal with the videodecoder capable of digital PSK demodulated signals and QAM demodulatedsignals.
 24. The signal receiving method in accordance with claim 23,wherein the decoding is MPEG decoding.